System and method for powering re-generation and re-transmission of millimeter waves for building penetration

ABSTRACT

A system enabling signal penetration into a building comprising first circuitry, located on an exterior of the building, for transmitting and receiving signals at a first frequency that experience losses when penetrating into an interior of the building, converting the received signals at the first frequency into a first format that overcome losses caused by penetrating into the interior of the building over a wireless communications link and converting received signals in the first format into the signals in the first frequency. A first antenna associated with the first circuitry transmits the signals in the first format into the interior of the building via a wireless communications link and receives signals from the interior of the building in the first format via the wireless communications link. First power circuitry provides system power to each of the first circuitry and the first antenna responsive to a provided power signal. Second circuitry, located on the interior of the building and communicatively linked with the first circuitry via the wireless communications link, for receives and transmits the converted received signals in the first format that counteracts the losses caused by penetrating into the interior of the building from/to the first circuitry. A second antenna associated with the second circuitry transmits the signals in the first format to the exterior of the building via the wireless communications link and for receives signals from the exterior of the building in the first format via the wireless communications link. Second power circuitry provides system power to each of the second circuitry and the second antenna responsive to a generated power signal. First wireless power transmission circuitry located on the interior of the building generates a wireless power signal for transmission to the exterior of the building over a wireless power link responsive to the provided power signal. Second wireless power transmission circuitry located on the exterior of the building receives the wireless power signal over the wireless power link and generates the generated power signal responsive to the wireless power signal.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.16/808,990, filed Mar. 4, 2020, entitled SYSTEM FOR MILLIMETER WAVEBUILDING PENETRATION USING BEAM FORMING AND BEAM STEERING, now U.S. Pat.No. 10,784,962, issued on Sep. 22, 2020, which is continuation of U.S.patent application Ser. No. 16/530,528, filed Aug. 2, 2019, entitledPATCH ANTENNA FOR WAVE AGILITY, now U.S. Pat. No. 10,778,332, issued onSep. 15, 2020, which is a continuation of U.S. patent application Ser.No. 15/926,087, filed Mar. 20, 2018, entitled RE-GENERATION ANDRE-TRANSMISSION OF MILLIMETER WAVES FOR BUILDING PENETRATION, issued asU.S. Pat. No. 10,374,710 on Aug. 6, 2019. U.S. patent application Ser.No. 15/926,087 claims benefit of U.S. Provisional Application No.62/474,937, filed Mar. 22, 2017, entitled PATCH ANTENNA FOR WAVEAGILITY, and claims benefit of U.S. Provisional Application No.62/549,314, filed Aug. 23, 2017, entitled 60 GHZ PRODUCT TO ENABLEMM-WAVE ACCESS INSIDE BUILDINGS, and claims benefit of U.S. ProvisionalApplication No. 62/550,219, filed Aug. 25, 2017, entitled WAVE AGILITYSYSTEM, and claims benefit of U.S. Provisional Application No.62/559,286, filed Sep. 15, 2017, entitled MILLIMETER WAVE BUILDINGPENETRATION SYSTEM FOR USE WITH COMBINED INTERNET, TV AND PHONE SERVICE,and claims benefit of U.S. Provisional Application No. 62/598,268, filedDec. 13, 2017, entitled MAGNETIC RESONANCE POWER TRANSFER, and U.S.Provisional Application No. 62/638,555, filed Mar. 5, 2018, entitledPON-FWA SYSTEM TO UTILIZE 5G CORE AND ACCESS WITH MILLIMETER WAVEPENETRATION SYSTEM OVER VLOTHA, each of which is incorporated herein byreference in its entirety.

U.S. patent application Ser. No. 15/926,087 is also aContinuation-In-Part of U.S. patent application Ser. No. 15/466,320,filed Mar. 22, 2017, entitled RE-GENERATION AND RE-TRANSMISSION OFMILLIMETER WAVES FOR BUILDING PENETRATION, which published as US2017-0195054 on Jul. 6, 2017, issued as U.S. Pat. No. 10,014,948 on Jul.3, 2018. U.S. patent application Ser. No. 15/466,320 claims benefit ofU.S. Provisional Application No. 62/317,829, filed Apr. 4, 2016,entitled RE-GENERATION AND RE-TRANSMISSION OF MILLIMETER WAVES FORBUILDING PENETRATION, and claims benefit of U.S. Provisional ApplicationNo. 62/321,245, filed Apr. 12, 2016, entitled RE-GENERATION ANDRE-TRANSMISSION OF MILLIMETER WAVES FOR BUILDING PENETRATION, and claimsbenefit of U.S. Provisional Application No. 62/368,417, filed Jul. 29,2016, entitled REGENERATION, RETRANSMISSION OF MILLIMETER WAVES FORINDOOR PENETRATION, and claims benefit of U.S. Provisional ApplicationNo. 62/369,393, filed Aug. 1, 2016, entitled REGENERATION,RETRANSMISSION OF MILLIMETER WAVES FOR INDOOR PENETRATION, and claimsbenefit of U.S. Provisional Application No. 62/425,432, filed Nov. 22,2016, entitled REGENERATION, RETRANSMISSION OF MILLIMETER WAVES FORBUILDING PENETRATION USING HORN ANTENNAS. U.S. patent application Ser.No. 15/466,320 is also a Continuation-In-Part of U.S. patent applicationSer. No. 15/357,808, filed Nov. 21, 2016, entitled SYSTEM AND METHOD FORCOMMUNICATION USING ORBITAL ANGULAR MOMENTUM WITH MULTIPLE LAYER OVERLAYMODULATION, issued as U.S. Pat. No. 9,712,238 on Jul. 18, 2017, which isa Continuation of U.S. patent application Ser. No. 15/144,297, filed May2, 2016, entitled SYSTEM AND METHOD FOR COMMUNICATION USING ORBITALANGULAR MOMENTUM WITH MULTIPLE LAYER OVERLAY MODULATION, issued as U.S.Pat. No. 9,503,258 on Nov. 22, 2016, which is a Continuation of U.S.patent application Ser. No. 14/323,082, filed Jul. 3, 2014, entitledSYSTEM AND METHOD FOR COMMUNICATION USING ORBITAL ANGULAR MOMENTUM WITHMULTIPLE LAYER OVERLAY MODULATION, issued as U.S. Pat. No. 9,331,875 onMay 3, 2016, which claims benefit of U.S. Provisional Application No.61/975,142, filed Apr. 4, 2014, entitled SYSTEM AND METHOD FORCOMMUNICATION USING ORBITAL ANGULAR MOMENTUM WITH MODULATION. U.S.application Ser. Nos. 16/808,990, 16/530,528, 15/926,087, 15/466,320;62/317,829; 62/321,245; 62/368,417; 62/369,393; 62/425,432; 15/357,808;15/144,297; 14/323,082; 61/975,142; U.S. Patent Publication No.2017-0195054; and U.S. Pat. Nos. 10,014,948, 9,712,238; 9,503,258; and9,331,875 are each incorporated herein by reference in their entireties.

U.S. patent application Ser. No. 15/926,087 is also aContinuation-In-Part of U.S. patent application Ser. No. 15/636,142,filed Jun. 28, 2017, entitled PATCH ANTENNA ARRAY FOR TRANSMISSION OFHERMITE-GAUSSIAN AND LAGUERRE GAUSSIAN BEAMS, issued as U.S. Pat. No.10,027,434 on Jul. 17, 2018, which is a Continuation of U.S. patentapplication Ser. No. 15/457,444, filed Mar. 13, 2017, entitled PATCHANTENNA ARRAY FOR TRANSMISSION OF HERMITE-GAUSSIAN AND LAGUERRE GAUSSIANBEAMS, issued as U.S. Pat. No. 9,793,615 on Oct. 17, 2017, which is aContinuation of U.S. patent application Ser. No. 15/187,315, filed Jun.20, 2016, entitled PATCH ANTENNA ARRAY FOR TRANSMISSION OFHERMITE-GAUSSIAN AND LAGUERRE GAUSSIAN BEAMS, issued as U.S. Pat. No.9,595,766 on Mar. 14, 2017. U.S. patent application Ser. No. 15/187,315claims benefit of U.S. Provisional Application No. 62/182,227, filedJun. 19, 2015, entitled PATCH ANTENNAS FOR TRANSMISSION OFHERMITE-GAUSSIAN AND LAGUARRE GAUSSIAN BEAMS, and claims benefit of U.S.Provisional Application No. 62/233,838, filed Sep. 28, 2015, entitledPATCH ANTENNAS FOR TRANSMISSION OF HERMITE-GAUSSIAN AND LAGUERREGAUSSIAN BEAMS, and claims benefit of U.S. Provisional Application No.62/242,056, filed Oct. 15, 2015, entitled METHOD FOR MANUFACTURING APATCH ANTENNA, and claims benefit of U.S. Provisional Application No.62/311,633, filed Mar. 22, 2016, entitled HYBRID PATCH ANTENNA WITHPARABOLIC REFLECTOR. U.S. application Ser. Nos. 15/926,087, 15/636,142;15/457,444; 15/187,315; 62/182,227; 62/233,838; 62/242,056; 62/311,633and U.S. Pat. Nos. 10,027,434, 9,793,615 and 9,595,766 are eachincorporated herein by reference in their entireties.

TECHNICAL FIELD

The present invention relates to millimeter wave transmissions, and moreparticularly, to a manner for improving building penetration formillimeter wave transmissions using beam forming and beam steering.

BACKGROUND

Millimeter wave transmissions were developed as a bandwidth plan formaking 1300 MHz of the local multipoint distribution service (LMDS)spectrum available within the United States. The millimeter wavetransmissions meet the needs for increased bandwidth availability due tothe increasing bandwidth and application requirements for wirelessmobile devices. However, while increasing bandwidth capabilities,millimeter wave transmissions have the problem of having very poorbuilding penetration capabilities. Signals are drastically degraded whenattempting to penetrate most building structures. This provides aserious problem since the vast majority of wireless signaling traffic isoriginated from within buildings and the inability to utilize millimeterwave bandwidths would drastically limit its implementation in the modernmarketplace. Thus, there is a need for some manner for improvingbuilding penetration characteristics of millimeter wave transmissions.

SUMMARY

The present invention, as disclosed and described herein, in one aspectthereof, comprises a system for enabling signal penetration into abuilding comprising first circuitry, located on an exterior of thebuilding, for transmitting and receiving signals at a first frequencythat experience losses when penetrating into an interior of thebuilding, converting the received signals at the first frequency into afirst format that overcome losses caused by penetrating into theinterior of the building over a wireless communications link andconverting received signals in the first format into the signals in thefirst frequency. A first antenna associated with the first circuitrytransmits the signals in the first format into the interior of thebuilding via a wireless communications link and receives signals fromthe interior of the building in the first format via the wirelesscommunications link. First power circuitry provides system power to eachof the first circuitry and the first antenna responsive to a providedpower signal. Second circuitry, located on the interior of the buildingand communicatively linked with the first circuitry via the wirelesscommunications link, for receives and transmits the converted receivedsignals in the first format that counteracts the losses caused bypenetrating into the interior of the building from/to the firstcircuitry. A second antenna associated with the second circuitrytransmits the signals in the first format to the exterior of thebuilding via the wireless communications link and for receives signalsfrom the exterior of the building in the first format via the wirelesscommunications link. Second power circuitry provides system power toeach of the second circuitry and the second antenna responsive to agenerated power signal. First wireless power transmission circuitrylocated on the interior of the building generates a wireless powersignal for transmission to the exterior of the building over a wirelesspower link responsive to the provided power signal. Second wirelesspower transmission circuitry located on the exterior of the buildingreceives the wireless power signal over the wireless power link andgenerates the generated power signal responsive to the wireless powersignal.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding, reference is now made to thefollowing description taken in conjunction with the accompanyingDrawings in which:

FIG. 1A is a block diagram of a building penetration system;

FIG. 1B illustrates the bi-directional nature of the buildingpenetration system for transmissions from the outside;

FIG. 1C illustrates the bi-directional nature of the buildingpenetration system for transmissions from the inside;

FIG. 1D illustrates a network deployment of the building penetrationsystem of FIG. 1A;

FIG. 2 illustrates millimeter wave transmissions between a base stationand receivers located both inside and outside of a building structure;

FIG. 3A illustrates a block diagram of an optical bridge fortransmitting millimeter wave transmissions through a window;

FIG. 3B illustrates a block diagram of an embodiment wherein receivedsignals are down converted to a level that more easily transmits througha window or wall;

FIG. 4 is a more detailed block diagram of the millimeter waveregeneration and retransmission circuitry;

FIG. 5 illustrates the misalignment losses associated with themillimeter wave regeneration and retransmission circuitry;

FIG. 6 illustrates the RF transceiver circuitry of the millimeter waveregeneration and retransmission circuitry;

FIG. 7 illustrates the optical focusing circuitry of the millimeter waveregeneration and retransmission circuitry;

FIG. 8 illustrates various techniques for increasing spectral efficiencywithin a transmitted signal;

FIG. 9 illustrates a particular technique for increasing spectralefficiency within a transmitted signal;

FIG. 10 illustrates a general overview of the manner for providingcommunication bandwidth between various communication protocolinterfaces;

FIG. 11 illustrates the manner for utilizing multiple level overlaymodulation with twisted pair/cable interfaces;

FIG. 12 illustrates a general block diagram for processing a pluralityof data streams within an optical communication system;

FIG. 13 is a functional block diagram of a system for generating orbitalangular momentum within a communication system;

FIG. 14 is a functional block diagram of the orbital angular momentumsignal processing block of FIG. 7;

FIG. 15 is a functional block diagram illustrating the manner forremoving orbital angular momentum from a received signal including aplurality of data streams;

FIG. 16 illustrates a single wavelength having two quanti-spinpolarizations providing an infinite number of signals having variousorbital angular momentums associated therewith;

FIG. 17A illustrates a plane wave having only variations in the spinangular momentum;

FIG. 17B illustrates a signal having both spin and orbital angularmomentum applied thereto;

FIGS. 18A-18C illustrate various signals having different orbitalangular momentum applied thereto;

FIG. 18D illustrates a propagation of Poynting vectors for various Eigenmodes;

FIG. 18E illustrates a spiral phase plate;

FIG. 19 illustrates a multiple level overlay modulation system;

FIG. 20 illustrates a multiple level overlay demodulator;

FIG. 21 illustrates a multiple level overlay transmitter system;

FIG. 22 illustrates a multiple level overlay receiver system;

FIGS. 23A-23K illustrate representative multiple level overlay signalsand their respective spectral power densities;

FIG. 24 illustrates comparisons of multiple level overlay signals withinthe time and frequency domain;

FIG. 25 illustrates a spectral alignment of multiple level overlaysignals for differing bandwidths of signals;

FIG. 26 illustrates an alternative spectral alignment of multiple leveloverlay signals;

FIG. 27 illustrates power spectral density for various signal layersusing a combined three layer multiple level overlay technique;

FIG. 28 illustrates power spectral density on a log scale for layersusing a combined three layer multiple level overlay modulation;

FIG. 29 illustrates a bandwidth efficiency comparison for square rootraised cosine versus multiple layer overlay for a symbol rate of 1/6;

FIG. 30 illustrates a bandwidth efficiency comparison between squareroot raised cosine and multiple layer overlay for a symbol rate of 1/4;

FIG. 31 illustrates a performance comparison between square root raisedcosine and multiple level overlay using ACLR;

FIG. 32 illustrates a performance comparison between square root raisedcosine and multiple lever overlay using out of band power;

FIG. 33 illustrates a performance comparison between square root raisedcosine and multiple lever overlay using band edge PSD;

FIG. 34 is a block diagram of a transmitter subsystem for use withmultiple level overlay;

FIG. 35 is a block diagram of a receiver subsystem using multiple leveloverlay;

FIG. 36 illustrates an equivalent discreet time orthogonal channel ofmodified multiple level overlay;

FIG. 37 illustrates the PSDs of multiple layer overlay, modifiedmultiple layer overlay and square root raised cosine;

FIG. 38 illustrates a bandwidth comparison based on −40 dBc out of bandpower bandwidth between multiple layer overlay and square root raisedcosine;

FIG. 39 illustrates equivalent discrete time parallel orthogonalchannels of modified multiple layer overlay;

FIG. 40 illustrates the channel power gain of the parallel orthogonalchannels of modified multiple layer overlay with three layers andT_(sym)=3;

FIG. 41 illustrates a spectral efficiency comparison based on ACLR1between modified multiple layer overlay and square root raised cosine;

FIG. 42 illustrates a spectral efficiency comparison between modifiedmultiple layer overlay and square root raised cosine based on OBP;

FIG. 43 illustrates a spectral efficiency comparison based on ACLR1between modified multiple layer overlay and square root raised cosine;

FIG. 44 illustrates a spectral efficiency comparison based on OBPbetween modified multiple layer overlay and square root raised cosine;

FIG. 45 illustrates a block diagram of a baseband transmitter for a lowpass equivalent modified multiple layer overlay system;

FIG. 46 illustrates a block diagram of a baseband receiver for a lowpass equivalent modified multiple layer overlay system;

FIG. 47 illustrates a free-space communication system;

FIG. 48 illustrates a block diagram of a free-space optics system usingorbital angular momentum and multi-level overlay modulation;

FIGS. 49A-49C illustrate the manner for multiplexing multiple datachannels into optical links to achieve higher data capacity;

FIG. 49D illustrates groups of concentric rings for a wavelength havingmultiple OAM valves;

FIG. 50 illustrates a WDM channel containing many orthogonal OAM beams;

FIG. 51 illustrates a node of a free-space optical system;

FIG. 52 illustrates a network of nodes within a free-space opticalsystem;

FIG. 53 illustrates a system for multiplexing between a free spacesignal and an RF signal;

FIG. 54 illustrates an embodiment using horn antennas for transmittingdata through a window or wall;

FIG. 55 illustrates a downlink losses in the embodiment of FIG. 54;

FIG. 56 illustrates up link signal strengths in the embodiment of FIG.54;

FIG. 57 illustrates up link signal strengths when a power amplifier islocated inside of the building in the embodiment of FIG. 54;

FIG. 58 illustrates gains and losses on a downlink of the embodiment ofFIG. 59 when no power amplifier is incorporated;

FIG. 59 illustrates signal strengths at various points of the uplinkwhen no power amplifier is provided in the embodiment of FIG. 54;

FIG. 60 illustrates shielding used incorporation with the embodiment ofFIG. 56;

FIG. 61 illustrates an embodiment using patch antennas for transmittingdata through a window or wall;

FIG. 62 illustrates a patch antenna array used in the embodiment of FIG.61;

FIG. 63 illustrates a patch antenna of a patch antenna array;

FIG. 64 illustrates an antenna gain simulation for a patch antenna;

FIG. 65 illustrates the generation of a highly directional, high gainbeam using a patch antenna array;

FIG. 66 illustrates a further embodiment of a microstrip patch antennaarray;

FIG. 67 illustrates a patch antenna element;

FIG. 68 illustrates the electronic radiating fields of a patch antenna;

FIG. 69 illustrates a top view of a multilayer patch antenna array;

FIG. 70 illustrates a side view of a multilayer patch antenna array;

FIG. 71 illustrates a first layer of a multilayer patch antenna array;

FIG. 72 illustrates a second layer of a multilayer patch antenna array;

FIG. 73 illustrates a transmitter for use with a multilayer patchantenna array;

FIG. 74 illustrates a multiplexed OAM signal transmitted from amultilayer patch antenna array;

FIG. 75 illustrates a receiver for use with a multilayer patch antennaarray;

FIG. 76 illustrates a 3-D model of a single rectangular patch antenna;

FIG. 77 illustrates the radiation pattern of the patch antenna of FIG.10;

FIG. 78a illustrates the radiation pattern of a circular array for anOAM mode order l=0;

FIG. 78b illustrates the radiation pattern for an OAM mode order l=0 inthe vicinity of the array axis;

FIG. 78c illustrates the radiation pattern for an OAM mode order l=1 inthe vicinity of the array axis;

FIG. 78d illustrates the radiation pattern for an OAM mode order l=2 inthe vicinity of the array axis;

FIG. 79 is a flow diagram illustrating the design and layout process ofa patch antenna;

FIG. 80 is a flow diagram illustrating the process for patterning acopper layer on a laminate for a patch antenna; and

FIG. 81 is a flow diagram illustrating a testing process for amanufactured patch antenna.

FIG. 82B illustrates an embodiment of a system for transmitting wirelesssignals through a window or wall using a Peraso chipset;

FIG. 83 illustrates the implementation of a repeater using a Perasochipset;

FIG. 84A is a top-level block diagram of a Peraso transceiver;

FIGS. 84B and 84C illustrate a detailed application diagram of a Perasochipset;

FIG. 82A illustrates an embodiment for transmitting RF signals through awindow or wall using an RF transceiver chipset;

FIG. 85 illustrates serial transmissions between Peraso transceivers;

FIG. 86 illustrates parallel transmissions between Peraso transceivers;

FIG. 87 is a functional block diagram of a Peraso transceiver located onan exterior window;

FIG. 88 illustrates a method for providing power to an external Perasotransmitter using a laser;

FIG. 89 illustrates alignment holes within a VCSEL;

FIG. 90 illustrates the use of alignment holes for aligning opticalcircuits of VCSELs;

FIG. 91 illustrates optical power coupling between VCSELs;

FIG. 92 illustrates a manner for powering external system componentsusing solar panels;

FIG. 93 illustrates a manner for powering external system componentsusing lasers;

FIG. 94 illustrates a manner for powering exterior components from aninterior power source using inductive coupling;

FIG. 95 illustrates a pair of circular loop's linked by mutualinductance;

FIG. 96 illustrates a table providing information relating toefficiencies of a coil;

FIG. 97 is a schematic diagram for coils coupled via inductive coupling;

FIG. 98 is a schematic diagram for coils coupled via magnetic resonance;

FIG. 99 illustrates a functional block diagram of a magnetic resonancewireless power transfer system;

FIG. 100 is a schematic diagram of magnetically coupled resonators;

FIG. 101 is a schematic diagram of a simple power generation circuit;

FIG. 102 schematically illustrates the use of impedance matching toovercome Eddy current losses;

FIG. 103 illustrates a perspective view of Peraso transceivercircuitries located on the exterior and interior of a structure;

FIG. 104 illustrates a side view of Peraso transceiver circuitrieslocated on an exterior and interior of a structure;

FIG. 105 illustrates a table of various parameters associated withtransmission of signals through window glass;

FIG. 106 illustrates another table of various parameters associated withtransmission of signals through window glass;

FIG. 107 illustrates a further table of various parameters associatedwith transmission of signals through window glass;

FIG. 108 illustrates the manner in which a millimeter wave system may becombined with a residential IP network for providing broadband datatransmission;

FIG. 109 illustrates a functional block diagram of a combined IPresidential network system;

FIG. 110 is a functional block diagram of a residential P networksystem;

FIG. 111 illustrates the manner in which a mmwave system may be utilizedto transmit information to a residential IP network system;

FIG. 112 illustrates a first embodiment for wireless transmission ofbroadband data to a residential IP network system;

FIG. 113 illustrates a second embodiment for wireless transmission ofbroadband data to a residential IP network system;

FIG. 114 illustrates a third embodiment for wireless transmission ofbroadband data to a residential IP network system;

FIG. 115 illustrates a combined optical data transfer system and RF datatransfer system for providing broadband data to a residential gateway;

FIG. 116 illustrates the manner in which load-balancing techniques maybe used to control network configuration between an optical network datatransfer system and an RF network data transfer system;

FIG. 117 illustrates various optical connections between a centraloffice and a customer premises;

FIG. 118 illustrates a GPON architecture;

FIG. 119 illustrates upstream and downstream GTC frames;

FIG. 120 illustrates a downstream GTC frame format;

FIG. 121 illustrates an upstream GTC frame format;

FIG. 122 illustrates a virtual optical line termination hardwareabstraction (vOLTHA) layer;

FIG. 123 illustrates an implementation of vOLTHA on an OLT and ONU link;

FIG. 124 illustrates a broadband link between an OLT and home gateway;

FIG. 125 illustrates the interface between and ONU and the plurality ofhome gateways;

FIG. 126 illustrates a first embodiment of a broadband datacommunications link between an OLT and home gateway;

FIG. 127 illustrates a second embodiment of a broadband datacommunications link between an OLT and virtual reality goggles;

FIG. 128 is a functional block diagram of a 60 GHz transceiver dongle;

FIG. 129 illustrates a six byte MAC address for an ethernet interfacewithin one of the above broadband communication links; and

FIG. 130 illustrates a switch within the PON network describedhereinabove.

DETAILED DESCRIPTION

Referring now to the drawings, wherein like reference numbers are usedherein to designate like elements throughout, the various views andembodiments of regeneration and retransmission of millimeter waves forbuilding penetration and various embodiments associated therewith areillustrated and described, and other possible embodiments are described.The figures are not necessarily drawn to scale, and in some instancesthe drawings have been exaggerated and/or simplified in places forillustrative purposes only. One of ordinary skill in the art willappreciate the many possible applications and variations based on thefollowing examples of possible embodiments.

One issue with wireless telecommunications is the inability of highfrequency RF waves to penetrate through windows and walls of homes andbusiness offices. If a window includes any infrared (IR) shielding inorder to save energy within the house or office building, the losses insignals transmitted through the shielding are typically up to 40 or 50dB. Thus, the millimeter wave system described herein provides theability to provide tunneling of such optical and high frequency radiowaves without requiring the need to drill through the glass, window orbuilding to provide a physical portal therethrough would provide greatbenefits to wireless communication technologies. This may be done at anyfrequency that has problems penetrating through the glass or building.Glass is one of the most popular and versatile due to its constantlyimproving solar and thermal performance. One manner for achieving thisperformance is through the use of passive and solar control lowemissivity coatings. These low emissivity glass materials produce a hugeloss for millimeter wave spectrum transmissions and create a hugeproblem for transmission of millimeter waves through such glass. Thesystem described herein below provides for the ability to allowfrequencies having a problem penetrating through a glass or building tobe processed in such a manner to enable the signals to be transmittedinto or out of a home or building.

Millimeter wave signaling was developed when the FCC devised a band planmaking 1300 MHz of the local multipoint distribution service (LMDS)spectrum available within each basic trading area across the UnitedStates. The plan allocated two LMDS licenses per BTA (basic tradingarea), an “A Block” and a “B Block” in each. The A Block licensecomprised 1150 MHz of total bandwidth, and the B Block license consistedof 150 MHz of total bandwidth. A license holder Teligent developed asystem for fixed wireless point to multipoint technology that could sendhigh speed broadband from rooftops to surrounding small and medium-sizebusinesses. However, the system, as well as others provided by Winstarand NextLink, did not succeed and many of the LMDS licenses fell backinto the hands of the FCC. These licenses and related spectrum are seenas useful for 5G trials and services.

Referring now to FIG. 1A, there is illustrated a general block diagramof the building penetration transmission system 102. The buildingpenetration transmission system 102 uses 5G fixed millimeter wavedeployments to overcome high building penetration losses due to RF andoptical obstructions such as windows, brick and concrete walls. Thebuilding penetration transmission system 102 greatly increases thenumber of enterprise and residential buildings where 5G millimeter wavesignals can be used to deliver gigabyte ethernet services. The systemprovides an optical or RF tunnel through the window or wall 106 withoutrequiring the drilling of any holes or the creation of some type ofsignal permeable portal within the window or wall. The generation ofdirection radio waves using the describe system enables the generationof directional beams to tunnel through low-e glass or walls. The systemenables link budgets between the interior and exterior transceviers besatisfied. The system greatly increases the number of building that mayuse millimeter wave signals to deliver Gigabit Ethernet using consumerinstalled devices.

The building penetration transmission system 102 generally includes anexterior repeater transmitter 104 located on the exterior of the windowor wall 106. The repeater transmitter 104 transmits and receives anumber of frequencies including 2.5 GHz band, 3.5 GHz band, 5 GHz band,24 GHz band, 28 GHz band (A1, A2, B1 and B2), 39 GHz band, 60 GHz band,71 GHz band and 81 GHz band. The 3.5 GHz band is CBRS (Citizens BandRadio Service), the 60 GHz band is V-band and the 71 GHz and 81 GHz areE-band. The repeater transmitter 104 is powered using magnetic resonanceor inductive coupling such that the outside unit requires no externalpower source. The repeater 104 transmits received signals through thewindow or wall 106 to a transceiver 108 located on the interior of thebuilding. The transceiver 108 includes an antenna 110 for providingethernet and/or power connections. The building penetration transmissionsystem 102 may provide one gigabit per second throughput traffictunneling through a building structure such as a window or wall. Thetransceiver 108 may include a port 112 providing femto cellconnectivity, but in general transmits Wi-Fi indoors using the antenna110. Alternatively, the ethernet or power connections can be hardwiredto the transceiver 108. The building penetration transmission system 102may be located at any point on a wall or window of a structure. Thebuilding penetration transmission system 102 is designed to work withdifferent types of walls and windows in order to enable millimeter wavesignals to penetrate different types of structures. The repeater 104 andtransceiver 108 are constructed of a metal/plastic design to withstandthe harshest environments including precipitation, hot/cold weather andhigh/low humidity.

The transceiver 108 includes gigabyte ethernet ports, a power output, atleast one USB 2.0 port and dual flash image support. The buildingpenetration transmission system 102 provides a range of up to 200 feet(60 m). The system requires a 24 V/M passive gigabyte PoE and has a 20 Wmaximum power consumption that may be powered using magnetic resonancewireless charging in one embodiment. The system provides 2 GHz ofchannel bandwidth 60 GHz.

FIGS. 1B and 1C illustrates the bidirectional communication betweentransceiver 104 located on the exterior side of the window or wall 106and transceiver 108 located on the interior side of the window or wall106. A remote base station transmitter 109 transmits wireless signals toan external transceiver 104. Communication transmissions from theexterior transceiver 104 to the interior transceiver 108 occur over acommunications link 114. The signals transmitted to the interior maythen be transmitted to consumer premises equipment (CPE) 111 using beamforming or WiFi 113 from an internal router 115. As shown in FIG. 1C,internal devices 117 (such as mobile devices or Internet-of-Thingsdevices) transmit signals to the internal router 115. The internalrouter 115 provides the signals to the internal transceiver 108.Transmissions from the interior of the window or wall 106 to theexterior are from transceiver 108 to transceiver 104 are oncommunications link 116. The external transceiver 104 then transmits thesignals to the external base station 109. Thus, the system enablesbidirectional communications that may utilize RF, optical or other typesof communication technologies as more fully described hereinbelow.

Referring now to FIG. 1D, there is illustrated a network deployment ofthe building penetration system discussed with respect to FIGS. 1A-1C. Aprovider network 130 interfaces with the local network through fiber PoP(point of presence) cabinets 132. The cabinets 132 have a fiber link 134to an access point 136. Each of the access points 136 wirelesslycommunicates with a network of other access points 136 that are locatedfor example on light poles within a local area over wirelesscommunication links using any number of communication frequencies aswill be described herein. The access points 136 communicate withtransceiver systems 138 that comprise the building penetration systemdescribed herein where in signals are wirelessly transmitted to anexternal transceiver and then transmitted to the interior of thebusiness or home such that information may be bi-directionallytransmitted from the provider network 130 to/from devices located withinthe interior of various structures. In this manner, data may be providedbetween the network provider 130 and devices of all types located withinthe structures using wireless communications that normally would notpenetrate to the interior of the structures due to losses occurring bypenetration of the signals into the interior of the structures.

Referring now to FIG. 2, there is illustrated the use of a millimeterwave transmission system 202 for communications. The base station 204generates the millimeter wave transmissions 206, 208 for transmissionsto various receivers 210, 212. Millimeter wave transmissions 206 thattraveled directly from the base station 204 to a receiver 210 are ableto be easily received without much ambient interference. Millimeter wavetransmissions 208 from a base station 204 to a receiver 212 locatedinside of the building 214 will have significant interference issues.Millimeter wave transmissions 208 do not easily penetrate a building204. When passing through transparent windows or building wallssignificant signal losses are experienced. The 28 GHz and abovefrequencies do not penetrate building walls and glass of the windows yet85% of communications traffic is generated from within buildings.

In view of millimeter wave spectrum transmissions not propagating veryfar and lacking the ability to penetrate indoors, these frequencies willbe used for very short range applications of about a mile. By way ofperspective, at 2.4 GHz, a low-power Wi-Fi can cover most of a housethat's under 3000 sq. ft., but a 5 GHz Wi-Fi signal would only coverapproximately 60% of a two-story house because the signal does nottravel as far at the higher frequency range. For 5G applications, thepower is higher, but still higher frequencies have higher losses andpropagation through space and other media.

The losses occurring as the millimeter wave signals penetrate a buildingdrive data rates down to almost nothing. For example, when transmittingon a downlink from a base station to the inside of a home or buildingthrough clear glass, the maximum data rate is 9.93 Gb per second. Whentransmitting through tinted glass the data rate is 2.2 Mb per second.When transmitting through brick the data rate is 14 Mb per second, andwhen transmitting through concrete, the data rate drops all the way to0.018 bps. Similarly, when transmitting on an uplink from the inside ofthe building towards a base station, the maximum data rate through clearglass is 1.57 Gb per second and through tinted glass is 0.37 Mb persecond. The signal being transmitted on the uplink has a data rate of5.5 Mb per second when transmitted through brick and 0.0075 bits persecond when transmitted through concrete. Differences are also providedon the downlink and uplink when transmitting to/from older or newerbuildings. Older buildings are defined as buildings using a compositemodel that comprises 30% standard glass and 70% concrete wall. Newerbuildings are defined as composite models comprising 70% infraredreflective glass (IRR glass) and 30% concrete wall. Base stationtransmissions on the downlink to the inside of the building are 32 Mbper second for older buildings and 0.32 Mb per second for newerbuildings. Similarly, the uplink transmissions from inside thehome/building to the base station are 2.56 Mb per second for olderbuildings in 25.6 kb per second for newer buildings.

Despite the shortcomings, in order to meet the increased demands forbandwidth, RF service providers will increasingly move to carrierfrequencies of higher frequency rates. In particular, 28 GHz is anemerging frequency band for providing local multipoint distributionservice (LMDS). The 28 GHz and 39 GHz frequency bands are beingcontemplated by the FCC for small cell deployments to support 5Gnetworks to subscriber premises using beam forming and beam steering.These higher frequency bandwidths have a number of advantages inaddition to the disadvantages caused by the huge penetration losses whenpassing through building materials or windows. These advantages includea higher frequency rate, capability of more precise beamforming and moreeffective beam steering in the smaller footprint of the componentsproviding the millimeter wave frequencies.

FIG. 3A illustrates one manner for transmitting millimeter wave signalsinside of a building using an optical bridge 302 mounted to a window304. The optical bridge 302 includes a first portion 306 included on anoutside of the window 304 and a second portion 308 included on theinside of the window 304. The first portion 306 includes a 28 GHztransceiver 310 that is mounted on the outside of the window 304. The 28GHz transceiver 310 receives the millimeter wave transmissions that arebeing transmitted from, for example, a base station 104 such as thatdescribed with respect to FIG. 1. The received/transmitted signals aretransmitted to and from the transceiver 310 using a receiver opticalsubassembly (ROSA)/transmission optical subassembly (TOSA) 312. Areceiver optical subassembly is a component used for receiving opticalsignals in a fiber optic system. Similarly, a transceiver opticalsubassembly is a component used for transmitting optical signals in afiber optic system. ROSA/TOSA component 312 transmits or receives theoptical signals through the window 304 to a ROSA/TOSA component 314located on the inside of the window 304. The signals are forwarded fromthe ROSA/TOSA 314 to a Wi-Fi transmitter 316 for transmissions withinthe building.

FIG. 3B illustrates a further embodiment wherein a received frequencythat does not easily penetrate a tinted window or wall 330 down convertsa received signal in order to facilitate transmission between the windowor wall 330. On the exterior of the building, a signal is received at anantenna 332 of a transceiver 334 at a frequency that does not easilypenetrate a window or wall. The transceiver 334 forwards the signals toa down/up converter 336 for down converting the signals to a frequencyband that will more easily penetrate the window/wall 330. Anothertransceiver 338 takes the frequency down converted signal from theconverter 336 and transmits it through the wall or window 330. Thetransmitted signal is received by a transceiver 340 located on theinterior of the building at the down converted frequency. The receivedsignal is passed to an up/down converter 342 to convert the signal to alevel for transmission in the interior of the building. In many casesthis may be the Wi-Fi band. The up converted signal is forwarded to arouter 344 for transmission within the building. Outgoing signalreceived from devices located within the building are processed andtransmitted in a reverse manner to transmit the signal outside of thebuilding from transceiver 334.

Referring now to FIG. 4, there is illustrated a more detailedillustration of the components for transmitting millimeter wavetransmissions through a window or wall of a building. The transceiver210 includes an optional antenna gain element 402 for receiving themillimeter wave transmissions transmitted on a down/up link 404 from abase station 204. The down/uplink 404 comprises a 28 GHz beamtransmission. However other frequency transmissions may also beutilized. An RF receiver 406 is used for receiving information from thebase station 204 over the down/up link 404. Similarly, the RFtransmitter 408 is used for transmitting information on the down/up link404 to a base station 204. Receive signals are provided to a demodulator410 for demodulation of any received signals. The demodulated signalsare provided to a groomer 412 which places the signals in theappropriate configuration for transmission by the optical transmissioncomponents. When translating different modulations (say from a highorder QAM to OOK (On-Off Keying)), there are signaling conversions thatrequire some grooming (or signal conditioning) to ensure all bitstranslate properly and still provide a low BER. The present systemtranslates from RF at a high QAM rate to raw bit rates of OOK to enabletransmissions using the VCSELs to go through the glass of the window.VCSELs only work with OOK and therefore a translation using the groomer412 is needed. If a received signal were just down-convert from 28 GHzdirectly to 5.8 GHz (because 5.8 GHz does pass through the wall andglass), then we do not need to worry about complications of translatingto low order modulation. The problem is that down-converting signal from28 GHz to 5.8 GHz requires expensive components. The groomer 412completes the translation of the received 28 GHz signal to a frequencyfor transmission through a glass or wall without the more expensivecomponents.

The signals to be transmitted are passed through an amplifier 414 toamplify the signal for transmission. The amplified signal is provided toVCSELs 416 for optically transmitting the signal. The VCSEL 416 is avertical cavity surface emitting laser that is a type of semiconductorlaser diode with laser beam omissions perpendicular from the topsurface. In a preferred embodiment, the VCSEL 416 comprises a FinisarVCSEL having a wavelength of approximately 780 nm, a modulation rate of4 Gb per second and an optical output power of 2.2 mW (3.4 to dBm). Inalternative embodiments the components for transmitting the opticalsignals across the window 404 may comprise an LED (light emitting diode)or EEL (edge emitting lasers). The different lasers enable differentoptical re-transmissions at different frequencies based on differentcharacteristics of a window such as tint.

The VCSEL 416 includes a transmission optical subassembly (TOSA) forgenerating the optical signals for transmission from VCSEL 416 to VCSEL418 located on the opposite side of the window 404. The VCSELs 416 and418 comprise a laser source for generating the optical signals fortransmission across the window 404. In one embodiment, the VCSELcomprises a Finisar VCSEL that provides a 780 nm optical signal having amaximum modulation rate of 4 Gb per second when running at 1 Gb persecond and an optical output power of 3 mW (5 dBm). The TOSA includes alaser device or LED device for converting electrical signals from theamplifier 414 into light signal transmissions. Transmissions from theoutside VCSEL 416 to the inside VCSEL 418 and an associated receiveroptical subassembly (ROSA).

The optical signals are transmitted through the window 404 using opticalfocusing circuitry 417. The optical focusing circuitry 417 will be morefully described on the transmitter and receiver sides with respect toFIG. 7. The optical link 428 between VCSEL 416 and VCSEL 418 has anoptical link budget associated therewith that defines the losses thatmay be accepted while still transmitting the information between theVCSELs 416, 418. The VCSEL has an output power of approximately 5 dBm.The detector at the receiver within the VCSEL can detect a signal atapproximately −12 dBm. The glass losses associated with the opticalsignal passing through the glass at a wavelength of 780 nm is 7.21 dB.The coupling loss and lens gain associated with the transmission isapproximately 0.1 dB. The maximum displacement loss caused by a lensdisplacement of 3.5 mm is 6.8 dB. Thus, the total link margin equals2.88 dB based upon a subtraction of the detector sensitivity, glasslosses, coupling loss and lens gain and maximum displacement loss fromthe VCSEL output power. The 2.88 dB link margin is provided forunexpected an extra losses such as len's losses and unexpected outputvariances.

Lens displacement or misalignment can account for a significant portionof the link loss within the system. As illustrated in FIG. 5, the rangeof tolerable misalignment 402 ranges from approximately −6.5 mm to +6.5mm from the center of the power spectrum received by the detector. Thealignment losses 404 range in an area from 0.6 dB to 6.8 dB as themisalignment moves between 6.5 mm. The maximum allowed misalignment lossis 9.4 dB as illustrated at 406.

The VCSEL 418 on the inside of the window 204 uses a TOSA to transmit anoptical signal at a data rate of 0.5 Gbps through the window 204 to aROSA within the VCSEL 416 located on the outside of the window. Thereceived optical signal is provided to a de-groomer component 32 forprocessing the signals from raw bit rates of OOK to RF at high QAM rateto enable RF transmissions after receipt of the signals by the VCSELs.The de-groomed signal is modulated within a modulator 422. The modulatedsignal is transmitted over the uplink 404 using an RF transmitter 408.The transceiver 310 is powered by a power input 424 the componentsinside the window are similarly powered by a power input 426. Signalsare provided within the building using a Wi-Fi transmitter 428 that isconnected to receive optical signals received by the VCSEL 418 andprovide signals to the VCSEL 418 for transmission through the window304. The Wi-Fi transmitter uses the 802.11 transmission protocol.

Referring now to FIG. 6 there is illustrated a more detailed blockdiagram of the transceiver 310. The receiver portion 602 includes an RFreceiver 604 for receiving the RF signals transmitted from the basestation on the downlink 606. The receiver 604 generates output signalshaving a real portion BBI 608 and an imaginary portion BBQ 610. The RFreceiver 604 generates the real signal 608 and imaginary signal 610responsive to the receive signal and inputs from a phase lockedloop/voltage control oscillator 605. The phase locked loop/voltagecontrol oscillator 605 provides inputs to the RF receiver 604 responsiveto a reference oscillator signal provided from reference oscillator 607and a voltage controlled oscillator signal provided from oscillator 609.The real signal 608 and the imaginary signal 610 are provided toanalog-to-digital converters 612 for conversion to a digital signal. Theanalog-to-digital converters 612 are clocked by an associated clockinput 614 provided from clock generation circuit 616. The clockgeneration circuit 616 also receives an input from the referenceoscillator 607. The real and imaginary digital signals 618 and 620 areinput to a digital down converter 622. The digital signals are downconverted to a lower frequency and output as a bit stream 624 to theoptical transmission circuitry (VCSEL) for transmitting across thewindow glass.

The transmitter portion 624 receives a digital bitstream 626 from theoptical circuitry and provides this bitstream to the real and imaginaryportions of digital up converters 628 to convert the digital data to ahigher frequency for transmission. The real and imaginary portions ofthe up-converted digital signal are provided to a crest factor reductionprocessor 630. Some signals (especially OFDM-based systems) have highpeak-to-average power ratio (PAR) that negatively impacts the efficiencyof power amplifiers (PAs). Crest factor reduction (CFR) schemesimplemented by the processor help reduce PAR and have been used for manynetworks (CDMA & OFDM). However, CFR schemes developed primarily forCDMA signals have a poor performance when used in OFDM (given the tighterror vector magnitude (EVM) requirements). With a well-designed CFRalgorithm on FPGAs, one can achieve low-latency, high-performance thatcan significantly reduce the PAR of the output signal which improves PAefficiency and reduced cost.

The real and imaginary signals are provided from the crest factorreduction processor 630 to a digital to analog converter 632. Thedigital to analog converter 632 converts the real and imaginary digitalsignals into real and imaginary analog signals BBI 634 and BBQ 636. Thereal and imaginary analog signals are inputs to the RF transmitter 638.The RF transmitter 638 processes the real signal 634 and imaginarysignal 636 responsive to input from the phase locked loop/voltagecontrol oscillator 604 to generate RF signals for transmission on theuplink 640 to generate the millimeter wave and transmissions.

Referring now to FIG. 7, there is illustrated the optical focusingcircuitry 317 associated with the optical transmission interface acrossthe window 304. The optical focusing circuitry 417 is included with theVCSEL located on each side of the window 204 and includes a transmissionportion 602 and a receiver portion 604. The transmission portion 602 andreceiver portion 604 would be included on each side of the window 304 asthe system provides bidirectional communications across the window. Thetransmission portion 602 includes in one embodiment a VCSEL 606 providedby Finisar that transmits a 780 nm optical signal at 4 Gb per second andhas a power output of 3.42 dBm. The optical signal generated by theVCSEL 606 is provided to an acromatic doublet 608 having a focal lengthof 7.5 mm that collimates the optical signal generated by the VCSEL 606into a small aperture. A collimated beam 610 is transmitted across thewindow 304. The collimated beam exits the window 304 and on the receiverportion 604 first passes through a bi-convex lens 612 having a focallength of 25 mm. The bi-convex lens 612 focuses the beam column 610 ontoa half ball lens 614 that focuses the optical signal onto asemiconductor aperture of a photo detector 616. In one embodiment, thedetector 616 has an aperture diameter of 10 mm and a detectorsensitivity of 12 dBm.

The transmissions between the VCSELs 606 and to and from the RFtransceiver to 10 may in one particular embodiment utilize orthogonalfunction signal transmission techniques such as those described in U.S.application Ser. No. 15/357,808, entitled SYSTEM AND METHOD FORCOMMUNICATION USING ORBITAL ANGULAR MOMENTUM WITH MULTIPLE LAYER OVERLAYMODULATION, filed on Nov. 21, 2016, which is incorporated herein byreference in its entirety. However, it should be realized that a varietyof other data transmission techniques may also be used.

FIG. 7 illustrates two manners for increasing spectral efficiency of acommunications system. In general, there are basically two ways toincrease spectral efficiency 702 of a communications system. Theincrease may be brought about by signal processing techniques 704 in themodulation scheme or using multiple access technique. Additionally, thespectral efficiency can be increase by creating new Eigen channels 706within the electromagnetic propagation. These two techniques arecompletely independent of one another and innovations from one class canbe added to innovations from the second class. Therefore, thecombination of this technique introduced a further innovation.

Spectral efficiency 702 is the key driver of the business model of acommunications system. The spectral efficiency is defined in units ofbit/sec/hz and the higher the spectral efficiency, the better thebusiness model. This is because spectral efficiency can translate to agreater number of users, higher throughput, higher quality or some ofeach within a communications system.

Regarding techniques using signal processing techniques or multipleaccess techniques. These techniques include innovations such as TDMA,FDMA, CDMA, EVDO, GSM, WCDMA, HSPA and the most recent OFDM techniquesused in 4G WIMAX and LTE. Almost all of these techniques use decades-oldmodulation techniques based on sinusoidal Eigen functions called QAMmodulation. Within the second class of techniques involving the creationof new Eigen channels 706, the innovations include diversity techniquesincluding space and polarization diversity as well as multipleinput/multiple output (MIMO) where uncorrelated radio paths createindependent Eigen channels and propagation of electromagnetic waves.

Referring now to FIG. 8, the communication system configurationintroduces two techniques, one from the signal processing techniques 804category and one from the creation of new eigen channels 806 categorythat are entirely independent from each other. Their combinationprovides a unique manner to disrupt the access part of an end to endcommunications system from twisted pair and cable to fiber optics, tofree space optics, to RF used in cellular, backhaul and satellite. Thefirst technique involves the use of a new signal processing techniqueusing new orthogonal signals to upgrade QAM modulation using nonsinusoidal functions. This particular embodiment is referred to asquantum level overlay (QLO) 902 as shown in FIG. 9. The secondembodiment involves the application of new electromagnetic wavefrontsusing a property of electromagnetic waves or photon, called orbitalangular momentum (QAM) 904. Application of each of the quantum leveloverlay techniques 902 and orbital angular momentum application 904uniquely offers orders of magnitude higher spectral efficiency 906within communication systems in their combination.

With respect to the quantum level overlay technique 902, new eigenfunctions are introduced that when overlapped (on top of one anotherwithin a symbol) significantly increases the spectral efficiency of thesystem. The quantum level overlay technique 302 borrows from quantummechanics, special orthogonal signals that reduce the time bandwidthproduct and thereby increase the spectral efficiency of the channel.Each orthogonal signal is overlaid within the symbol acts as anindependent channel. These independent channels differentiate thetechnique from existing modulation techniques.

With respect to the application of orbital angular momentum 904, thisembodiment introduces twisted electromagnetic waves, or light beams,having helical wave fronts that carry orbital angular momentum (OAM).Different OAM carrying waves/beams can be mutually orthogonal to eachother within the spatial domain, allowing the waves/beams to beefficiently multiplexed and demultiplexed within a communications link.OAM beams are interesting in communications due to their potentialability in special multiplexing multiple independent data carryingchannels.

With respect to the combination of quantum level overlay techniques 902and orbital angular momentum application 904, the combination is uniqueas the OAM multiplexing technique is compatible with otherelectromagnetic techniques such as wave length and polarization divisionmultiplexing. This suggests the possibility of further increasing systemperformance. The application of these techniques together in highcapacity data transmission disrupts the access part of an end to endcommunications system from twisted pair and cable to fiber optics, tofree space optics, to RF used in cellular/backhaul and satellites.

Each of these techniques can be applied independent of one another, butthe combination provides a unique opportunity to not only increasespectral efficiency, but to increase spectral efficiency withoutsacrificing distance or signal to noise ratios.

Using the Shannon Capacity Equation, a determination may be made ifspectral efficiency is increased. This can be mathematically translatedto more bandwidth. Since bandwidth has a value, one can easily convertspectral efficiency gains to financial gains for the business impact ofusing higher spectral efficiency. Also, when sophisticated forward errorcorrection (FEC) techniques are used, the net impact is higher qualitybut with the sacrifice of some bandwidth. However, if one can achievehigher spectral efficiency (or more virtual bandwidth), one cansacrifice some of the gained bandwidth for FEC and therefore higherspectral efficiency can also translate to higher quality.

Telecom operators and vendors are interested in increasing spectralefficiency. However, the issue with respect to this increase is thecost. Each technique at different layers of the protocol has a differentprice tag associated therewith. Techniques that are implemented at aphysical layer have the most impact as other techniques can besuperimposed on top of the lower layer techniques and thus increase thespectral efficiency further. The price tag for some of the techniquescan be drastic when one considers other associated costs. For example,the multiple input multiple output (MIMO) technique uses additionalantennas to create additional paths where each RF path can be treated asan independent channel and thus increase the aggregate spectralefficiency. In the MIMO scenario, the operator has other associated softcosts dealing with structural issues such as antenna installations, etc.These techniques not only have tremendous cost, but they have hugetiming issues as the structural activities take time and the achievingof higher spectral efficiency comes with significant delays which canalso be translated to financial losses.

The quantum level overlay technique 902 has an advantage that theindependent channels are created within the symbols without needing newantennas. This will have a tremendous cost and time benefit compared toother techniques. Also, the quantum layer overlay technique 902 is aphysical layer technique, which means there are other techniques athigher layers of the protocol that can all ride on top of the QLOtechniques 902 and thus increase the spectral efficiency even further.QLO technique 902 uses standard QAM modulation used in OFDM basedmultiple access technologies such as WIMAX or LTE. QLO technique 902basically enhances the QAM modulation at the transceiver by injectingnew signals to the I & Q components of the baseband and overlaying thembefore QAM modulation as will be more fully described herein below. Atthe receiver, the reverse procedure is used to separate the overlaidsignal and the net effect is a pulse shaping that allows betterlocalization of the spectrum compared to standard QAM or even the rootraised cosine. The impact of this technique is a significantly higherspectral efficiency.

Referring now more particularly to FIG. 10, there is illustrated ageneral overview of the manner for providing improved communicationbandwidth within various communication protocol interfaces 1002, using acombination of multiple level overlay modulation 1004 and theapplication of orbital angular momentum 1006 to increase the number ofcommunications channels. The following discussions of orbital angularmomentum processing and multiple level overlay modulation illustrate twotechniques that may or may not be implemented in RF transmissions in thebelow described systems and embodiments. RF transmissions may beconfigured to implement one, both or neither of the techniques in thedescribed embodiments.

The various communication protocol interfaces 1002 may comprise avariety of communication links, such as RF communication, wirelinecommunication such as cable or twisted pair connections, or opticalcommunications making use of light wavelengths such as fiber-opticcommunications or free-space optics. Various types of RF communicationsmay include a combination of RF microwave or RF satellite communication,as well as multiplexing between RF and free-space optics in real time.

By combining a multiple layer overlay modulation technique 1004 withorbital angular momentum (OAM) technique 1006, a higher throughput overvarious types of communication links 1002 may be achieved. The use ofmultiple level overlay modulation alone without OAM increases thespectral efficiency of communication links 1002, whether wired, optical,or wireless. However, with OAM, the increase in spectral efficiency iseven more significant.

Multiple overlay modulation techniques 1004 provide a new degree offreedom beyond the conventional 2 degrees of freedom, with time T andfrequency F being independent variables in a two-dimensional notationalspace defining orthogonal axes in an information diagram. This comprisesa more general approach rather than modeling signals as fixed in eitherthe frequency or time domain. Previous modeling methods using fixed timeor fixed frequency are considered to be more limiting cases of thegeneral approach of using multiple level overlay modulation 1004. Withinthe multiple level overlay modulation technique 1004, signals may bedifferentiated in two-dimensional space rather than along a single axis.Thus, the information-carrying capacity of a communications channel maybe determined by a number of signals which occupy different time andfrequency coordinates and may be differentiated in a notationaltwo-dimensional space.

Within the notational two-dimensional space, minimization of the timebandwidth product, i.e., the area occupied by a signal in that space,enables denser packing, and thus, the use of more signals, with higherresulting information-carrying capacity, within an allocated channel.Given the frequency channel delta (Δf), a given signal transmittedthrough it in minimum time Δt will have an envelope described by certaintime-bandwidth minimizing signals. The time-bandwidth products for thesesignals take the form;ΔtΔf=½(2n+1)where n is an integer ranging from 0 to infinity, denoting the order ofthe signal.

These signals form an orthogonal set of infinite elements, where eachhas a finite amount of energy. They are finite in both the time domainand the frequency domain, and can be detected from a mix of othersignals and noise through correlation, for example, by match filtering.Unlike other wavelets, these orthogonal signals have similar time andfrequency forms.

The orbital angular momentum process 1006 provides a twist to wavefronts of the electromagnetic fields carrying the data stream that mayenable the transmission of multiple data streams on the same frequency,wavelength, or other signal-supporting mechanism. This will increase thebandwidth over a communications link by allowing a single frequency orwavelength to support multiple eigen channels, each of the individualchannels having a different orthogonal and independent orbital angularmomentum associated therewith.

Referring now to FIG. 11, there is illustrated a further communicationimplementation technique using the above described techniques as twistedpairs or cables carry electrons (not photons). Rather than using each ofthe multiple level overlay modulation 1004 and orbital angular momentumtechniques 1006, only the multiple level overlay modulation 1004 can beused in conjunction with a single wireline interface and, moreparticularly, a twisted pair communication link or a cable communicationlink 1102. The operation of the multiple level overlay modulation 1104,is similar to that discussed previously with respect to FIG. 10, but isused by itself without the use of orbital angular momentum techniques1006, and is used with either a twisted pair communication link or cableinterface communication link 1102.

Referring now to FIG. 12, there is illustrated a general block diagramfor processing a plurality of data streams 1202 for transmission in anoptical communication system. The multiple data streams 1202 areprovided to the multi-layer overlay modulation circuitry 1204 whereinthe signals are modulated using the multi-layer overlay modulationtechnique. The modulated signals are provided to orbital angularmomentum processing circuitry 1206 which applies a twist to each of thewave fronts being transmitted on the wavelengths of the opticalcommunication channel. The twisted waves are transmitted through theoptical interface 1208 over an optical communications link such as anoptical fiber or free space optics communication system. FIG. 12 mayalso illustrate an RF mechanism wherein the interface 1208 wouldcomprise and RF interface rather than an optical interface.

Referring now more particularly to FIG. 13, there is illustrated afunctional block diagram of a system for generating the orbital angularmomentum “twist” within a communication system, such as that illustratedwith respect to FIG. 10, to provide a data stream that may be combinedwith multiple other data streams for transmission upon a same wavelengthor frequency. Multiple data streams 1302 are provided to thetransmission processing circuitry 1300. Each of the data streams 1302comprises, for example, an end to end link connection carrying a voicecall or a packet connection transmitting non-circuit switch packed dataover a data connection. The multiple data streams 1302 are processed bymodulator/demodulator circuitry 1304. The modulator/demodulatorcircuitry 1304 modulates the received data stream 1302 onto a wavelengthor frequency channel using a multiple level overlay modulationtechnique, as will be more fully described herein below. Thecommunications link may comprise an optical fiber link, free-spaceoptics link, RF microwave link, RF satellite link, wired link (withoutthe twist), etc.

The modulated data stream is provided to the orbital angular momentum(OAM) signal processing block 1306. Each of the modulated data streamsfrom the modulator/demodulator 1304 are provided a different orbitalangular momentum by the orbital angular momentum electromagnetic block1306 such that each of the modulated data streams have a unique anddifferent orbital angular momentum associated therewith. Each of themodulated signals having an associated orbital angular momentum areprovided to an optical transmitter 1308 that transmits each of themodulated data streams having a unique orbital angular momentum on asame wavelength. Each wavelength has a selected number of bandwidthslots B and may have its data transmission capability increase by afactor of the number of degrees of orbital angular momentum l that areprovided from the OAM electromagnetic block 1306. The opticaltransmitter 1308 transmitting signals at a single wavelength couldtransmit B groups of information. The optical transmitter 1308 and OAMelectromagnetic block 1306 may transmit l×B groups of informationaccording to the configuration described herein.

In a receiving mode, the optical transmitter 1308 will have a wavelengthincluding multiple signals transmitted therein having different orbitalangular momentum signals embedded therein. The optical transmitter 1308forwards these signals to the OAM signal processing block 1306, whichseparates each of the signals having different orbital angular momentumand provides the separated signals to the demodulator circuitry 1304.The demodulation process extracts the data streams 1302 from themodulated signals and provides it at the receiving end using themultiple layer overlay demodulation technique.

Referring now to FIG. 14, there is provided a more detailed functionaldescription of the OAM signal processing block 1406. Each of the inputdata streams are provided to OAM circuitry 1402. Each of the OAMcircuitry 1402 provides a different orbital angular momentum to thereceived data stream. The different orbital angular momentums areachieved by applying different currents for the generation of thesignals that are being transmitted to create a particular orbitalangular momentum associated therewith. The orbital angular momentumprovided by each of the OAM circuitries 1402 are unique to the datastream that is provided thereto. An infinite number of orbital angularmomentums may be applied to different input data streams using manydifferent currents. Each of the separately generated data streams areprovided to a signal combiner 1404, which combines the signals onto awavelength for transmission from the transmitter 1406.

Referring now to FIG. 15, there is illustrated the manner in which theOAM processing circuitry 1306 may separate a received signal intomultiple data streams. The receiver 1502 receives the combined OAMsignals on a single wavelength and provides this information to a signalseparator 1504. The signal separator 1504 separates each of the signalshaving different orbital angular momentums from the received wavelengthand provides the separated signals to OAM de-twisting circuitry 1506.The OAM de-twisting circuitry 1506 removes the associated OAM twist fromeach of the associated signals and provides the received modulated datastream for further processing. The signal separator 1504 separates eachof the received signals that have had the orbital angular momentumremoved therefrom into individual received signals. The individuallyreceived signals are provided to the receiver 1502 for demodulationusing, for example, multiple level overlay demodulation as will be morefully described herein below.

FIG. 16 illustrates in a manner in which a single wavelength orfrequency, having two quanti-spin polarizations may provide an infinitenumber of twists having various orbital angular momentums associatedtherewith. The l axis represents the various quantized orbital angularmomentum states which may be applied to a particular signal at aselected frequency or wavelength. The symbol omega (o) represents thevarious frequencies to which the signals of differing orbital angularmomentum may be applied. The top grid 1602 represents the potentiallyavailable signals for a left handed signal polarization, while thebottom grid 1604 is for potentially available signals having righthanded polarization.

By applying different orbital angular momentum states to a signal at aparticular frequency or wavelength, a potentially infinite number ofstates may be provided at the frequency or wavelength. Thus, the stateat the frequency Δω or wavelength 1606 in both the left handedpolarization plane 1602 and the right handed polarization plane 1604 canprovide an infinite number of signals at different orbital angularmomentum states Δl. Blocks 1608 and 1610 represent a particular signalhaving an orbital angular momentum Δl at a frequency Δω or wavelength inboth the right handed polarization plane 1604 and left handedpolarization plane 1610, respectively. By changing to a differentorbital angular momentum within the same frequency Δω or wavelength1606, different signals may also be transmitted. Each angular momentumstate corresponds to a different determined current level fortransmission from the optical transmitter. By estimating the equivalentcurrent for generating a particular orbital angular momentum within theoptical domain and applying this current for transmission of thesignals, the transmission of the signal may be achieved at a desiredorbital angular momentum state.

Thus, the illustration of FIG. 16, illustrates two possible angularmomentums, the spin angular momentum, and the orbital angular momentum.The spin version is manifested within the polarizations of macroscopicelectromagnetism, and has only left and right hand polarizations due toup and down spin directions. However, the orbital angular momentumindicates an infinite number of states that are quantized. The paths aremore than two and can theoretically be infinite through the quantizedorbital angular momentum levels.

Using the orbital angular momentum state of the transmitted energysignals, physical information can be embedded within the radiationtransmitted by the signals. The Maxwell-Heaviside equations can berepresented as:

${\nabla{\cdot E}} = \frac{\rho}{ɛ_{0}}$${\nabla{\times E}} = {- \frac{\partial B}{\partial t}}$ ∇⋅B = 0${\nabla{\times B}} = {{ɛ_{0}\mu_{0}\frac{\partial E}{\partial t}} + {\mu_{0}{j\left( {t,x} \right)}}}$where ∇ is the del operator, E is the electric field intensity and B isthe magnetic flux density. Using these equations, one can derive 23symmetries/conserved quantities from Maxwell's original equations.However, there are only ten well-known conserved quantities and only afew of these are commercially used. Historically if Maxwell's equationswhere kept in their original quaternion forms, it would have been easierto see the symmetries/conserved quantities, but when they were modifiedto their present vectorial form by Heaviside, it became more difficultto see such inherent symmetries in Maxwell's equations.

Maxwell's linear theory is of U(1) symmetry with Abelian commutationrelations. They can be extended to higher symmetry group SU(2) form withnon-Abelian commutation relations that address global (non-local inspace) properties. The Wu-Yang and Harmuth interpretation of Maxwell'stheory implicates the existence of magnetic monopoles and magneticcharges. As far as the classical fields are concerned, these theoreticalconstructs are pseudo-particle, or instanton. The interpretation ofMaxwell's work actually departs in a significant ways from Maxwell'soriginal intention. In Maxwell's original formulation, Faraday'selectrotonic states (the A field) was central making them compatiblewith Yang-Mills theory (prior to Heaviside). The mathematical dynamicentities called solitons can be either classical or quantum, linear ornon-linear and describe EM waves. However, solitons are of SU(2)symmetry forms. In order for conventional interpreted classicalMaxwell's theory of U(1) symmetry to describe such entities, the theorymust be extended to SU(2) forms.

Besides the half dozen physical phenomena (that cannot be explained withconventional Maxwell's theory), the recently formulated Harmuth Ansatzalso address the incompleteness of Maxwell's theory. Harmuth amendedMaxwell's equations can be used to calculate EM signal velocitiesprovided that a magnetic current density and magnetic charge are addedwhich is consistent to Yang-Mills filed equations. Therefore, with thecorrect geometry and topology, the A potentials always have physicalmeaning

The conserved quantities and the electromagnetic field can berepresented according to the conservation of system energy and theconservation of system linear momentum. Time symmetry, i.e. theconservation of system energy can be represented using Poynting'stheorem according to the equations:

$\begin{matrix}{H = {{\sum\limits_{i}{m_{i}\gamma_{i}c^{2}}} + {\frac{ɛ_{0}}{2}{\int{d^{3} \times \left( {{E}^{2} + {c^{2}{B}^{2}}} \right)}}}}} & {{Hamiltonian}\mspace{14mu}\left( {{total}\mspace{14mu}{energy}} \right)} \\{\mspace{79mu}{{\frac{d\; U^{mech}}{dt} + \frac{d\; U^{em}}{dt} + {\oint_{s^{\prime}}{d^{2}x^{\prime}{\hat{n^{\prime}} \cdot S}}}} = 0}} & {{conservation}\mspace{14mu}{of}\mspace{14mu}{energy}}\end{matrix}$

The space symmetry, i.e., the conservation of system linear momentumrepresenting the electromagnetic Doppler shift can be represented by theequations:

$\begin{matrix}{\mspace{79mu}{p = {{\sum\limits_{i}{m_{i}\gamma_{i}v_{i}}} + {ɛ_{0}{\int{d^{3}{x\left( {E \times B} \right)}}}}}}} & {{linear}\mspace{14mu}{momentum}} \\{{\frac{{dp}^{mech}}{dt} + \frac{{dp}^{em}}{dt} + {\oint_{s^{\prime}}{d^{2}x^{\prime}{\hat{n^{\prime}} \cdot T}}}} = 0} & {{conservation}\mspace{14mu}{of}\mspace{14mu}{linear}\mspace{14mu}{momentum}}\end{matrix}$The conservation of system center of energy is represented by theequation:

$R = {{\frac{1}{H}{\sum\limits_{i}{\left( {x_{i} - x_{0}} \right)m_{i}\gamma_{i}c^{2}}}} + {\frac{ɛ_{0}}{2\; H}{\int{d^{3}{x\left( {x - x_{0}} \right)}\left( {{E^{2}} + {c^{2}{B^{2}}}} \right)}}}}$

Similarly, the conservation of system angular momentum, which gives riseto the azimuthal Doppler shift is represented by the equation:

$\begin{matrix}{{\frac{{dJ}^{mech}}{dt} + \frac{{dJ}^{em}}{dt} + {\oint_{s^{\prime}}{d^{2}x^{\prime}{\hat{n^{\prime}} \cdot M}}}} = 0} & {{conservation}\mspace{14mu}{of}\mspace{14mu}{angular}\mspace{14mu}{momentum}}\end{matrix}$For radiation beams in free space, the EM field angular momentum J^(em)can be separated into two parts:J ^(em)=ε₀∫_(V′) d ³ x′(E×A)+ε₀∫_(V′) d ³ x′E _(i)[(x′−x ₀)×∇]A _(i)For each singular Fourier mode in real valued representation:

$J^{em} = {{{- i}\;\frac{ɛ_{0}}{2\;\omega}{\int_{V^{\prime}}{d^{3}{x^{\prime}\left( {E^{*} \times E} \right)}}}} - {i\;\frac{ɛ_{0}}{2\;\omega}{\int_{V^{\prime}}{d^{3}x^{\prime}{E_{i}\left\lbrack {\left( {x^{\prime} - x_{0}} \right) \times \nabla} \right\rbrack}E_{i}}}}}$

The first part is the EM spin angular momentum S^(em), its classicalmanifestation is wave polarization. And the second part is the EMorbital angular momentum L^(em) its classical manifestation is wavehelicity. In general, both EM linear momentum P^(em), and EM angularmomentum J^(em)=L^(em)+S^(em) are radiated all the way to the far field.

By using Poynting theorem, the optical vorticity of the signals may bedetermined according to the optical velocity equation:

$\begin{matrix}{{{\frac{\partial U}{\partial t} + {\nabla{\cdot S}}} = 0},} & {{continuity}\mspace{14mu}{equation}}\end{matrix}$where S is the Poynting vectorS=¼(E×H*+E*×H),and U is the energy densityU=¼(ε|E| ²+μ₀ |H| ²),with E and H comprising the electric field and the magnetic field,respectively, and ε and μ₀ being the permittivity and the permeabilityof the medium, respectively. The optical vorticity V may then bedetermined by the curl of the optical velocity according to theequation:

$V = {{\nabla{\times v_{opt}}} = {\nabla{\times \left( \frac{{E \times H^{*}} + {E^{*} \times H}}{{ɛ{E}^{2}} + {\mu_{0}{H}^{2}}} \right)}}}$

Referring now to FIGS. 17A and 17B, there is illustrated the manner inwhich a signal and its associated Poynting vector in a plane wavesituation. In the plane wave situation illustrated generally at 1702,the transmitted signal may take one of three configurations. When theelectric field vectors are in the same direction, a linear signal isprovided, as illustrated generally at 1704. Within a circularpolarization 1706, the electric field vectors rotate with the samemagnitude. Within the elliptical polarization 1708, the electric fieldvectors rotate but have differing magnitudes. The Poynting vectorremains in a constant direction for the signal configuration to FIG. 17Aand always perpendicular to the electric and magnetic fields. Referringnow to FIG. 17B, when a unique orbital angular momentum is applied to asignal as described here and above, the Poynting vector S 1710 willspiral about the direction of propagation of the signal. This spiral maybe varied in order to enable signals to be transmitted on the samefrequency as described herein.

FIGS. 18A through 18C illustrate the differences in signals havingdifferent helicity (i.e., orbital angular momentums). Each of thespiraling Poynting vectors associated with the signals 1802, 1804, and1806 provide a different shaped signal. Signal 1802 has an orbitalangular momentum of +1, signal 1804 has an orbital angular momentum of+3, and signal 1806 has an orbital angular momentum of −4. Each signalhas a distinct angular momentum and associated Poynting vector enablingthe signal to be distinguished from other signals within a samefrequency. This allows differing type of information to be transmittedon the same frequency, since these signals are separately detectable anddo not interfere with each other (Eigen channels).

FIG. 18D illustrates the propagation of Poynting vectors for variousEigen modes. Each of the rings 1820 represents a different Eigen mode ortwist representing a different orbital angular momentum within the samefrequency. Each of these rings 1820 represents a different orthogonalchannel. Each of the Eigen modes has a Poynting vector 1822 associatedtherewith.

Topological charge may be multiplexed to the frequency for either linearor circular polarization. In case of linear polarizations, topologicalcharge would be multiplexed on vertical and horizontal polarization. Incase of circular polarization, topological charge would multiplex onleft hand and right hand circular polarizations. The topological chargeis another name for the helicity index “I” or the amount of twist or OAMapplied to the signal. The helicity index may be positive or negative.In RF, different topological charges can be created and muxed togetherand de-muxed to separate the topological charges.

The topological charges e s can be created using Spiral Phase Plates(SPPs) as shown in FIG. 18E using a proper material with specific indexof refraction and ability to machine shop or phase mask, hologramscreated of new materials or a new technique to create an RF version ofSpatial Light Modulator (SLM) that does the twist of the RF waves (asopposed to optical beams) by adjusting voltages on the device resultingin twisting of the RF waves with a specific topological charge. SpiralPhase plates can transform a RF plane wave (

=0) to a twisted RF wave of a specific helicity (i.e.

=+1).

Cross talk and multipath interference can be corrected using RFMultiple-Input-Multiple-Output (MIMO). Most of the channel impairmentscan be detected using a control or pilot channel and be corrected usingalgorithmic techniques (closed loop control system).

As described previously with respect to FIG. 13, each of the multipledata streams applied within the processing circuitry has a multiplelayer overlay modulation scheme applied thereto.

Referring now to FIG. 19, the reference number 1900 generally indicatesan embodiment of a multiple level overlay (MLO) modulation system,although it should be understood that the term MLO and the illustratedsystem 1900 are examples of embodiments. The MLO system may comprise onesuch as that disclosed in U.S. Pat. No. 8,503,546 entitled MultipleLayer Overlay Modulation which is incorporated herein by reference. Inone example, the modulation system 1900 would be implemented within themultiple level overlay modulation box 1204 of FIG. 12. System 1900 takesas input an input data stream 1901 from a digital source 1902, which isseparated into three parallel, separate data streams, 1903A-1903C, oflogical is and Os by input stage demultiplexer (DEMUX) 1904. Data stream1901 may represent a data file to be transferred, or an audio or videodata stream. It should be understood that a greater or lesser number ofseparated data streams may be used. In some of the embodiments, each ofthe separated data streams 1903A-1903C has a data rate of 1/N of theoriginal rate, where N is the number of parallel data streams. In theembodiment illustrated in FIG. 19, N is 3.

Each of the separated data streams 1903A-1903C is mapped to a quadratureamplitude modulation (QAM) symbol in an M-QAM constellation, forexample, 16 QAM or 64 QAM, by one of the QAM symbol mappers 1905A-C. TheQAM symbol mappers 1905A-C are coupled to respective outputs of DEMUX1904, and produced parallel in phase (I) 1906A, 1908A, and 1910A andquadrature phase (Q) 1906B, 1908B, and 1910B data streams at discretelevels. For example, in 64 QAM, each I and Q channel uses 8 discretelevels to transmit 3 bits per symbol. Each of the three I and Q pairs,1906A-1906B, 1908A-1908B, and 1910A-1910B, is used to weight the outputof the corresponding pair of function generators 1907A-1907B,1909A-1909B, and 1911A-1911B, which in some embodiments generate signalssuch as the modified Hermite polynomials described above and weightsthem based on the amplitude value of the input symbols. This provides 2Nweighted or modulated signals, each carrying a portion of the dataoriginally from income data stream 1901, and is in place of modulatingeach symbol in the I and Q pairs, 1906A-1906B, 1908A-1908B, and1910A-1910B with a raised cosine filter, as would be done for a priorart QAM system. In the illustrated embodiment, three signals are used,SH0, SH1, and SH2, which correspond to modifications of H0, H1, and H2,respectively, although it should be understood that different signalsmay be used in other embodiments.

The weighted signals are not subcarriers, but rather are sublayers of amodulated carrier, and are combined, superimposed in both frequency andtime, using summers 1912 and 1916, without mutual interference in eachof the I and Q dimensions, due to the signal orthogonality. Summers 1912and 1916 act as signal combiners to produce composite signals 1913 and1917. The weighted orthogonal signals are used for both I and Qchannels, which have been processed equivalently by system 1900, and aresummed before the QAM signal is transmitted. Therefore, although neworthogonal functions are used, some embodiments additionally use QAM fortransmission. Because of the tapering of the signals in the time domain,as will be shown in FIGS. 23A through 23K, the time domain waveform ofthe weighted signals will be confined to the duration of the symbols.Further, because of the tapering of the special signals and frequencydomain, the signal will also be confined to frequency domain, minimizinginterface with signals and adjacent channels.

The composite signals 1913 and 1917 are converted to analogue signals1915 and 1919 using digital to analogue converters 1914 and 1918, andare then used to modulate a carrier signal at the frequency of localoscillator (LO) 1920, using modulator 1921. Modulator 1921 comprisesmixers 1922 and 1924 coupled to DACs 1914 and 1918, respectively. Ninetydegree phase shifter 1923 converts the signals from LO 1920 into a Qcomponent of the carrier signal. The output of mixers 1922 and 1924 aresummed in summer 1925 to produce output signals 1926.

MLO can be used with a variety of transport mediums, such as wire,optical, and wireless, and may be used in conjunction with QAM. This isbecause MLO uses spectral overlay of various signals, rather thanspectral overlap. Bandwidth utilization efficiency may be increased byan order of magnitude, through extensions of available spectralresources into multiple layers. The number of orthogonal signals isincreased from 2, cosine and sine, in the prior art, to a number limitedby the accuracy and jitter limits of generators used to produce theorthogonal polynomials. In this manner, MLO extends each of the I and Qdimensions of QAM to any multiple access techniques such as GSM, codedivision multiple access (CDMA), wide band CDMA (WCDMA), high speeddownlink packet access (HSPDA), evolution-data optimized (EV-DO),orthogonal frequency division multiplexing (OFDM), world-wideinteroperability for microwave access (WIMAX), and long term evolution(LTE) systems. MLO may be further used in conjunction with othermultiple access (MA) schemes such as frequency division duplexing (FDD),time division duplexing (TDD), frequency division multiple access(FDMA), and time division multiple access (TDMA). Overlaying individualorthogonal signals over the same frequency band allows creation of avirtual bandwidth wider than the physical bandwidth, thus adding a newdimension to signal processing. This modulation is applicable to twistedpair, cable, fiber optic, satellite, broadcast, free-space optics, andall types of wireless access. The method and system are compatible withmany current and future multiple access systems, including EV-DO, UMB,WIMAX, WCDMA (with or without), multimedia broadcast multicast service(MBMS)/multiple input multiple output (MIMO), HSPA evolution, and LTE.

Referring now to FIG. 20, an MLO demodulator 2000 is illustrated,although it should be understood that the term MLO and the illustratedsystem 2000 are examples of embodiments. The demodulator 2000 takes asinput an MLO signal 2026 which may be similar to output signal 1926 fromsystem 1900. Synchronizer 2027 extracts phase information, which isinput to local oscillator 2020 to maintain coherence so that themodulator 2021 can produce base band to analogue I signal 2015 and Qsignal 2019. The modulator 2021 comprises mixers 2022 and 2024, which,coupled to LO 2020 through 90 degree phase shifter 2023. I signal 2015is input to each of signal filters 2007A, 2009A, and 2011A, and Q signal2019 is input to each of signal filters 2007B, 2009B, and 2011B. Sincethe orthogonal functions are known, they can be separated usingcorrelation or other techniques to recover the modulated data.Information in each of the I and Q signals 2015 and 2019 can beextracted from the overlapped functions which have been summed withineach of the symbols because the functions are orthogonal in acorrelative sense.

In some embodiments, signal filters 2007A-2007B, 2009A-2009B, and2011A-2011B use locally generated replicas of the polynomials as knownsignals in match filters. The outputs of the match filters are therecovered data bits, for example, equivalence of the QAM symbols1906A-1906B, 1908A-1908B, and 1910A-1910B of system 1900. Signal filters2007A-2007B, 2009A-2009B, and 2011A-2011B produce 2n streams of n, I,and Q signal pairs, which are input into demodulators 2028-2033.Demodulators 2028-2033 integrate the energy in their respective inputsignals to determine the value of the QAM symbol, and hence the logicalis and Os data bit stream segment represented by the determined symbol.The outputs of the demodulators 2028-2033 are then input intomultiplexers (MUXs) 2005A-2005C to generate data streams 2003A-2003C. Ifsystem 2000 is demodulating a signal from system 1900, data streams2003A-2003C correspond to data streams 1903A-1903C. Data streams2003A-2003C are multiplexed by MUX 2004 to generate data output stream2001. In summary, MLO signals are overlayed (stacked) on top of oneanother on transmitter and separated on receiver.

MLO may be differentiated from CDMA or OFDM by the manner in whichorthogonality among signals is achieved. MLO signals are mutuallyorthogonal in both time and frequency domains, and can be overlaid inthe same symbol time bandwidth product. Orthogonality is attained by thecorrelation properties, for example, by least sum of squares, of theoverlaid signals. In comparison, CDMA uses orthogonal interleaving ordisplacement of signals in the time domain, whereas OFDM uses orthogonaldisplacement of signals in the frequency domain.

Bandwidth efficiency may be increased for a channel by assigning thesame channel to multiple users. This is feasible if individual userinformation is mapped to special orthogonal functions. CDMA systemsoverlap multiple user information and views time intersymbol orthogonalcode sequences to distinguish individual users, and OFDM assigns uniquesignals to each user, but which are not overlaid, are only orthogonal inthe frequency domain. Neither CDMA nor OFDM increases bandwidthefficiency. CDMA uses more bandwidth than is necessary to transmit datawhen the signal has a low signal to noise ratio (SNR). OFDM spreads dataover many subcarriers to achieve superior performance in multipathradiofrequency environments. OFDM uses a cyclic prefix OFDM to mitigatemultipath effects and a guard time to minimize intersymbol interference(ISI), and each channel is mechanistically made to behave as if thetransmitted waveform is orthogonal. (Sync function for each subcarrierin frequency domain.)

In contrast, MLO uses a set of functions which effectively form analphabet that provides more usable channels in the same bandwidth,thereby enabling high bandwidth efficiency. Some embodiments of MLO donot require the use of cyclic prefixes or guard times, and therefore,outperforms OFDM in spectral efficiency, peak to average power ratio,power consumption, and requires fewer operations per bit. In addition,embodiments of MLO are more tolerant of amplifier nonlinearities thanare CDMA and OFDM systems.

FIG. 21 illustrates an embodiment of an MLO transmitter system 2100,which receives input data stream 1901. System 2100 represents amodulator/controller 2101, which incorporates equivalent functionalityof DEMUX 1904, QAM symbol mappers 1905A-C, function generators1907A-1907B, 1909A-1909B, and 1911A-1911B, and summers 1912 and 1916 ofsystem 1900, shown in FIG. 19. However, it should be understood thatmodulator/controller 2101 may use a greater or lesser quantity ofsignals than the three illustrated in system 1900. Modulator/controller2101 may comprise an application specific integrated circuit (ASIC), afield programmable gate array (FPGA), and/or other components, whetherdiscrete circuit elements or integrated into a single integrated circuit(IC) chip.

Modulator/controller 2101 is coupled to DACs 2104 and 2107,communicating a 10 bit I signal 2102 and a 10 bit Q signal 2105,respectively. In some embodiments, I signal 2102 and Q signal 2105correspond to composite signals 1913 and 1917 of system 1900. It shouldbe understood, however, that the 10 bit capacity of I signal 2102 and Qsignal 2105 is merely representative of an embodiment. As illustrated,modulator/controller 2101 also controls DACs 2104 and 2107 using controlsignals 2103 and 2106, respectively. In some embodiments, DACs 2104 and2107 each comprise an AD5433, complementary metal oxide semiconductor(CMOS) 10 bit current output DAC. In some embodiments, multiple controlsignals are sent to each of DACs 2104 and 2107.

DACs 2104 and 2107 output analogue signals 2115 and 2119 to quadraturemodulator 1921, which is coupled to LO 1920. The output of modulator1921 is illustrated as coupled to a transmitter 2108 to transmit datawirelessly, although in some embodiments, modulator 1921 may be coupledto a fiber-optic modem, a twisted pair, a coaxial cable, or othersuitable transmission media.

FIG. 22 illustrates an embodiment of an MLO receiver system 2200 capableof receiving and demodulating signals from system 2100. System 2200receives an input signal from a receiver 2208 that may comprise inputmedium, such as RF, wired or optical. The modulator 2021 driven by LO2020 converts the input to baseband I signal 2015 and Q signal 2019. Isignal 2015 and Q signal 2019 are input to analogue to digital converter(ADC) 2209.

ADC 2209 outputs 10 bit signal 2210 to demodulator/controller 2201 andreceives a control signal 2212 from demodulator/controller 2201.Demodulator/controller 2201 may comprise an application specificintegrated circuit (ASIC), a field programmable gate array (FPGA),and/or other components, whether discrete circuit elements or integratedinto a single integrated circuit (IC) chip. Demodulator/controller 2201correlates received signals with locally generated replicas of thesignal set used, in order to perform demodulation and identify thesymbols sent. Demodulator/controller 2201 also estimates frequencyerrors and recovers the data clock, which is used to read data from theADC 2209. The clock timing is sent back to ADC 2209 using control signal2212, enabling ADC 2209 to segment the digital I and Q signals 2015 and2019. In some embodiments, multiple control signals are sent bydemodulator/controller 2201 to ADC 2209. Demodulator/controller 2201also outputs data signal 2001.

Hermite polynomials are a classical orthogonal polynomial sequence,which are the Eigenstates of a quantum harmonic oscillator. Signalsbased on Hermite polynomials possess the minimal time-bandwidth productproperty described above, and may be used for embodiments of MLOsystems. However, it should be understood that other signals may also beused, for example orthogonal polynomials such as Jacobi polynomials,Gegenbauer polynomials, Legendre polynomials, Chebyshev polynomials, andLaguerre polynomials. Q-functions are another class of functions thatcan be employed as a basis for MLO signals.

In quantum mechanics, a coherent state is a state of a quantum harmonicoscillator whose dynamics most closely resemble the oscillating behaviorof a classical harmonic oscillator system. A squeezed coherent state isany state of the quantum mechanical Hilbert space, such that theuncertainty principle is saturated. That is, the product of thecorresponding two operators takes on its minimum value. In embodimentsof an MLO system, operators correspond to time and frequency domainswherein the time-bandwidth product of the signals is minimized. Thesqueezing property of the signals allows scaling in time and frequencydomain simultaneously, without losing mutual orthogonality among thesignals in each layer. This property enables flexible implementations ofMLO systems in various communications systems.

Because signals with different orders are mutually orthogonal, they canbe overlaid to increase the spectral efficiency of a communicationchannel. For example, when n=0, the optimal baseband signal will have atime-bandwidth product of 1/2, which is the Nyquist Inter-SymbolInterference (ISI) criteria for avoiding ISI. However, signals withtime-bandwidth products of 3/2, 5/2, 7/2, and higher, can be overlaid toincrease spectral efficiency.

An embodiment of an MLO system uses functions based on modified Hermitepolynomials, 4n, and are defined by:

${\psi_{n}\left( {t,\xi} \right)} = {\frac{\left( {\tanh\;\xi} \right)^{n\text{/}2}}{2^{n\text{/}2}\left( {{n!}\cosh\;\xi} \right)^{1\text{/}2}}e^{\frac{1}{2}{t^{2}{\lbrack{1 - {\tanh\;\xi}}\rbrack}}}{H_{n}\left( \frac{t}{\sqrt{2\;\cosh\;\xi\;\sinh\;\xi}} \right)}}$where t is time, and ξ is a bandwidth utilization parameter. Plots ofΨ_(n) for n ranging from 0 to 9, along with their Fourier transforms(amplitude squared), are shown in FIGS. 5A-5K. The orthogonality ofdifferent orders of the functions may be verified by integrating:∫∫ψ_(n)(t,ξ)ψ_(m)(t,ξ)dtdξThe Hermite polynomial is defined by the contour integral:

${{H_{n}(z)} = {\frac{n!}{2\;\pi\; i}{\oint{e^{{- t^{2}} + {2\; t\; 2}}t^{{- n} - 1}{dt}}}}},$where the contour encloses the origin and is traversed in acounterclockwise direction. Hermite polynomials are described inMathematical Methods for Physicists, by George Arfken, for example onpage 416, the disclosure of which is incorporated by reference.

FIGS. 23A-23K illustrate representative MLO signals and their respectivespectral power densities based on the modified Hermite polynomials Ψ_(n)for n ranging from 0 to 9. FIG. 23A shows plots 2301 and 2304. Plot 2301comprises a curve 2327 representing Ψ₀ plotted against a time axis 2302and an amplitude axis 2303. As can be seen in plot 2301, curve 2327approximates a Gaussian curve. Plot 2304 comprises a curve 2337representing the power spectrum of Ψ₀ plotted against a frequency axis2305 and a power axis 2306. As can be seen in plot 2304, curve 2337 alsoapproximates a Gaussian curve. Frequency domain curve 2307 is generatedusing a Fourier transform of time domain curve 2327. The units of timeand frequency on axis 2302 and 2305 are normalized for basebandanalysis, although it should be understood that since the time andfrequency units are related by the Fourier transform, a desired time orfrequency span in one domain dictates the units of the correspondingcurve in the other domain. For example, various embodiments of MLOsystems may communicate using symbol rates in the megahertz (MHz) orgigahertz (GHz) ranges and the non-0 duration of a symbol represented bycurve 2327, i.e., the time period at which curve 2327 is above 0 wouldbe compressed to the appropriate length calculated using the inverse ofthe desired symbol rate. For an available bandwidth in the megahertzrange, the non-0 duration of a time domain signal will be in themicrosecond range.

FIGS. 23B-23J show plots 2307-2324, with time domain curves 2328-2336representing Ψ₁ through Ψ₉, respectively, and their correspondingfrequency domain curves 2338-2346. As can be seen in FIGS. 23A-23J, thenumber of peaks in the time domain plots, whether positive or negative,corresponds to the number of peaks in the corresponding frequency domainplot. For example, in plot 2323 of FIG. 23J, time domain curve 2336 hasfive positive and five negative peaks. In corresponding plot 2324therefore, frequency domain curve 2346 has ten peaks.

FIG. 23K shows overlay plots 2325 and 2326, which overlay curves2327-2336 and 2337-2346, respectively. As indicated in plot 2325, thevarious time domain curves have different durations. However, in someembodiments, the non-zero durations of the time domain curves are ofsimilar lengths. For an MLO system, the number of signals usedrepresents the number of overlays and the improvement in spectralefficiency. It should be understood that, while ten signals aredisclosed in FIGS. 23A-23K, a greater or lesser quantity of signals maybe used, and that further, a different set of signals, rather than the %n signals plotted, may be used.

MLO signals used in a modulation layer have minimum time-bandwidthproducts, which enable improvements in spectral efficiency, and arequadratically integrable. This is accomplished by overlaying multipledemultiplexed parallel data streams, transmitting them simultaneouslywithin the same bandwidth. The key to successful separation of theoverlaid data streams at the receiver is that the signals used withineach symbols period are mutually orthogonal. MLO overlays orthogonalsignals within a single symbol period. This orthogonality prevents ISIand inter-carrier interference (ICI).

Because MLO works in the baseband layer of signal processing, and someembodiments use QAM architecture, conventional wireless techniques foroptimizing air interface, or wireless segments, to other layers of theprotocol stack will also work with MLO. Techniques such as channeldiversity, equalization, error correction coding, spread spectrum,interleaving and space-time encoding are applicable to MLO. For example,time diversity using a multipath-mitigating rake receiver can also beused with MLO. MLO provides an alternative for higher order QAM, whenchannel conditions are only suitable for low order QAM, such as infading channels. MLO can also be used with CDMA to extend the number oforthogonal channels by overcoming the Walsh code limitation of CDMA. MLOcan also be applied to each tone in an OFDM signal to increase thespectral efficiency of the OFDM systems.

Embodiments of MLO systems amplitude modulate a symbol envelope tocreate sub-envelopes, rather than sub-carriers. For data encoding, eachsub-envelope is independently modulated according to N-QAM, resulting ineach sub-envelope independently carrying information, unlike OFDM.Rather than spreading information over many sub-carriers, as is done inOFDM, for MLO, each sub-envelope of the carrier carries separateinformation. This information can be recovered due to the orthogonalityof the sub-envelopes defined with respect to the sum of squares overtheir duration and/or spectrum. Pulse train synchronization or temporalcode synchronization, as needed for CDMA, is not an issue, because MLOis transparent beyond the symbol level. MLO addresses modification ofthe symbol, but since CDMA and TDMA are spreading techniques of multiplesymbol sequences over time. MLO can be used along with CDMA and TDMA.

FIG. 24 illustrates a comparison of MLO signal widths in the time andfrequency domains. Time domain envelope representations 2401-2403 ofsignals SH0-SH3 are illustrated as all having a duration Ts. SH0-SH3 mayrepresent PSI₀-PSI₂, or may be other signals. The correspondingfrequency domain envelope representations are 2405-2407, respectively.SH0 has a bandwidth BW, SH1 has a bandwidth three times BW, and SH2 hasa bandwidth of 5BW, which is five times as great as that of SH0. Thebandwidth used by an MLO system will be determined, at least in part, bythe widest bandwidth of any of the signals used. If each layer uses onlya single signal type within identical time windows, the spectrum willnot be fully utilized, because the lower order signals will use less ofthe available bandwidth than is used by the higher order signals.

FIG. 25 illustrates a spectral alignment of MLO signals that accountsfor the differing bandwidths of the signals, and makes spectral usagemore uniform, using SH0-SH3. Blocks 2501-2504 are frequency domainblocks of an OFDM signal with multiple subcarriers. Block 2503 isexpanded to show further detail. Block 2503 comprises a first layer 2503x comprised of multiple SH0 envelopes 2503 a-2503 o. A second layer 2503y of SH1 envelopes 2503 p-2503 t has one third the number of envelopesas the first layer. In the illustrated example, first layer 2503 x has15 SH0 envelopes, and second layer 2503 y has five SH1 envelopes. Thisis because, since the SH1 bandwidth envelope is three times as wide asthat of SH0, 15 SH0 envelopes occupy the same spectral width as five SH1envelopes. The third layer 2503 z of block 2503 comprises three SH2envelopes 2503 u-2503 w, because the SH2 envelope is five times thewidth of the SH0 envelope.

The total required bandwidth for such an implementation is a multiple ofthe least common multiple of the bandwidths of the MLO signals. In theillustrated example, the least common multiple of the bandwidth requiredfor SH0, SH1, and SH2 is 15BW, which is a block in the frequency domain.The OFDM-MLO signal can have multiple blocks, and the spectralefficiency of this illustrated implementation is proportional to(15+5+3)/15.

FIG. 26 illustrates another spectral alignment of MLO signals, which maybe used alternatively to alignment scheme shown in FIG. 25. In theembodiment illustrated in FIG. 26, the OFDM-MLO implementation stacksthe spectrum of SH0, SH1, and SH2 in such a way that the spectrum ineach layer is utilized uniformly. Layer 2600A comprises envelopes2601A-2601D, which includes both SH0 and SH2 envelopes. Similarly, layer2600C, comprising envelopes 2603A-2603D, includes both SH0 and SH2envelopes. Layer 2600B, however, comprising envelopes 2602A-2602D,includes only SH1 envelopes. Using the ratio of envelope sizes describedabove, it can be easily seen that BW+5BW=3BW+3BW. Thus, for each SH0envelope in layer 2600A, there is one SH2 envelope also in layer 2600Cand two SH1 envelopes in layer 2600B.

Three Scenarios Compared:

1) MLO with 3 Layers defined by:

${{f_{0}(t)} = {W_{0}e^{- \frac{t^{2}}{4}}}},{W_{0} = 0.6316}$${{f_{1}(t)} = {W_{1}{te}^{- \frac{t^{2}}{4}}}},{W_{1} \approx 0.6316}$${{f_{2}(t)} = {{W_{2}\left( {t^{2} - 1} \right)}e^{- \frac{t^{2}}{4}}}},{W_{2} \approx 0.4466}$(The current FPGA implementation uses the truncation interval of [−6,6].)2) Conventional scheme using rectangular pulse3) Conventional scheme using a square-root raised cosine (SRRC) pulsewith a roll-off factor of 0.5

For MLO pulses and SRRC pulse, the truncation interval is denoted by[−t1, t1] in the following figures. For simplicity, we used the MLOpulses defined above, which can be easily scaled in time to get thedesired time interval (say micro-seconds or nano-seconds). For the SRRCpulse, we fix the truncation interval of [−3T, 3T] where T is the symbolduration for all results presented in this document.

Bandwidth Efficiency

The X-dB bounded power spectral density bandwidth is defined as thesmallest frequency interval outside which the power spectral density(PSD) is X dB below the maximum value of the PSD. The X-dB can beconsidered as the out-of-band attenuation.

The bandwidth efficiency is expressed in Symbols per second per Hertz.The bit per second per Hertz can be obtained by multiplying the symbolsper second per Hertz with the number of bits per symbol (i.e.,multiplying with log 2 M for M-ary QAM).

Truncation of MLO pulses introduces inter-layer interferences (ILI).However, the truncation interval of [−6, 6] yields negligible ILI while[−4, 4] causes slight tolerable ILI.

The bandwidth efficiency of MLO may be enhanced by allowing inter-symbolinterference (ISI). To realize this enhancement, designing transmitterside parameters as well as developing receiver side detection algorithmsand error performance evaluation can be performed.

Referring now to FIG. 27, there is illustrated the power spectraldensity of each layer SH0-SH2 within MLO and also for the combined threelayer MLO. 2702 illustrates the power spectral density of the SH0 layer;2704 illustrates the power spectral density of the SH1 layer; 2706illustrates the power spectral density of the SH2 layer, and 2708illustrates the combined power spectral density of each layer.

Referring now to FIG. 28, there is illustrated the power spectraldensity of each layer as well as the power spectral density of thecombined three layer in a log scale. 2802 represents the SH0 layer. 2804represents the SH1 layer. 2806 represents the SH2 layer. 2808 representsthe combined layers.

Referring now to FIG. 29, there is a bandwidth efficiency comparisonversus out of band attenuation (X-dB) where quantum level overlay pulsetruncation interval is [−6,6] and the symbol rate is 1/6. Referring alsoto FIG. 30, there is illustrated the bandwidth efficiency comparisonversus out of band attenuation (X-dB) where quantum level overlay pulsetruncation interval is [−6,6] and the symbol rate is 1/4.

The QLO signals are generated from the Physicist's special Hermitefunctions:

${{f_{n}\left( {t,\alpha} \right)} = {\sqrt{\frac{\alpha}{\sqrt{\pi}{n!}2^{n}}}{H_{n}\left( {\alpha\; t} \right)}e^{- \frac{\alpha^{2}t^{2}}{2}}}},{\alpha > 0}$Note that the initial hardware implementation is using

$\alpha = \frac{1}{\sqrt{2}}$and for consistency with his part,

$\alpha = \frac{1}{\sqrt{2}}$is used in all figures related to the spectral efficiency.

Let the low-pass-equivalent power spectral density (PSD) of the combinedQLO signals be X(f) and its bandwidth be B. Here the bandwidth isdefined by one of the following criteria. ACLR1 (First Adjacent ChannelLeakage Ratio) in dBc equals:

${{ACLR}\; 1} = \frac{\int_{B\text{/}2}^{3\; B\text{/}2}{{X(f)}{df}}}{\int_{- \infty}^{\infty}{{X(f)}{df}}}$ACLR2 (Second Adjacent Channel Leakage Ratio) in dBc equals:

${{ACLR}\; 2} = \frac{\int_{3\; B\text{/}2}^{5\; B\text{/}2}{{X(f)}{df}}}{\int_{- \infty}^{\infty}{{X(f)}{df}}}$Out-of-Band Power to Total Power Ratio is:

$\frac{2\;{\int_{B\text{/}2}^{\infty}{{X(f)}{df}}}}{\int_{- \infty}^{\infty}{{X(f)}{df}}}$The Band-Edge PSD in dBc/100 kHz equals:

$\frac{\int_{B\text{/}2}^{\frac{B}{2} + 10^{5}}{{X(f)}{df}}}{\int_{- \infty}^{\infty}{{X(f)}{df}}}$

Referring now to FIG. 31 there is illustrated a performance comparisonusing ACLR1 and ACLR2 for both a square root raised cosine scheme and amultiple layer overlay scheme. Line 3102 illustrates the performance ofa square root raised cosine 3102 using ACLR1 versus an MLO 3104 usingACLR1. Additionally, a comparison between a square root raised cosine3106 using ACLR2 versus MLO 3108 using ACLR2 is illustrated. Table Aillustrates the performance comparison using ACLR.

TABLE A Criteria. ACLR1 ≤ −30 dBc per bandwidth Spectral EfficiencyACLR2 ≤ −43 dBc per bandwidth (Symbol/sec/Hz) Gain SRRC [−8T, 8T] β =0.22 0.8765 1.0 Symbol Duration N Layers (Tmol) QLO N = 3 Tmol = 4 1.1331.2926 [−8, 8] N = 4 Tmol = 5 1.094 1.2481 Tmol = 4 1.367 1.5596 N = 10Tmol = 8 1.185 1.3520 Tmol = 7 1.355 1.5459 Tmol = 6 1.580 1.8026 Tmol =5 1.896 2.1631 Tmol = 4 2.371 2.7051

Referring now to FIG. 32, there is illustrated a performance comparisonbetween a square root raised cosine 3202 and a MLO 3204 usingout-of-band power. Referring now also to Table B, there is illustrated amore detailed comparison of the performance using out-of-band power.

TABLE B Table 3: Performance Comparison Using Out-of-Band PowerCriterion: Spectral Efficiency Out-of-band Power: Total Power ≤ −30 dB(Symbol/sec/Hz) Gain SRRC [−8T, 8T] β = 0.22 0.861 1.0 Symbol Duration NLayers (Tmol) QLO N = 3 Tmol = 4 1.080 1.2544 [−8, 8] N = 4 Tmol = 51.049 1.2184 Tmol = 4 1.311 1.5226 N = 10 Tmol = 8 1.152 1.3380 Tmol = 71.317 1.5296 Tmol = 6 1.536 1.7840 Tmol = 5 1.844 2.1417 Tmol = 4 2.3052.6771

Referring now to FIG. 33, there is further provided a performancecomparison between a square root raised cosine 3302 and a MLO 3304 usingband-edge PSD. A more detailed illustration of the performancecomparison is provided in Table C.

TABLE C Table 4: Performance Comparison Using Band-Edge PSD Criterion:Spectral Efficiency Band-Edge PSD = −50 dBc/100 kHz (Symbol/sec/Hz) GainSRRC [−8T, 8T] β = 0.22 0.810 1.0 Symbol Duration N Layers (Tmol) QLO N= 3 Tmol = 4 0.925 1.1420 [−8, 8] N = 4 Tmol = 5 0.912 1.1259 Tmol = 41.14 1.4074 N = 10 Tmol = 8 1.049 1.2951 Tmol = 7 1.198 1.4790 Tmol = 61.398 1.7259 Tmol = 5 1.678 2.0716 Tmol = 4 2.097 2.5889

Referring now to FIGS. 34 and 35, there are more particularlyillustrated the transmit subsystem (FIG. 34) and the receiver subsystem(FIG. 35). The transceiver is realized using basic building blocksavailable as Commercially Off The Shelf products. Modulation,demodulation and Special Hermite correlation and de-correlation areimplemented on a FPGA board. The FPGA board 3402 at the receiver 3400estimated the frequency error and recovers the data clock (as well asdata), which is used to read data from the analog-to-digital (ADC) board3406. The FGBA board 3400 also segments the digital I and Q channels.

On the transmitter side 3400, the FPGA board 3402 realizes the specialhermite correlated QAM signal as well as the necessary control signalsto control the digital-to-analog (DAC) boards 3404 to produce analog I&Qbaseband channels for the subsequent up conversion within the directconversion quad modulator 3406. The direct conversion quad modulator3406 receives an oscillator signal from oscillator 3408.

The ADC 3506 receives the I&Q signals from the quad demodulator 3508that receives an oscillator signal from 3510.

Neither power amplifier in the transmitter nor an LNA in the receiver isused since the communication will take place over a short distance. Thefrequency band of 2.4-2.5 GHz (ISM band) is selected, but any frequencyband of interest may be utilized.

MIMO uses diversity to achieve some incremental spectral efficiency.Each of the signals from the antennas acts as an independent orthogonalchannel. With QLO, the gain in spectral efficiency comes from within thesymbol and each QLO signal acts as independent channels as they are allorthogonal to one another in any permutation. However, since QLO isimplemented at the bottom of the protocol stack (physical layer), anytechnologies at higher levels of the protocol (i.e. Transport) will workwith QLO. Therefore one can use all the conventional techniques withQLO. This includes RAKE receivers and equalizers to combat fading,cyclical prefix insertion to combat time dispersion and all othertechniques using beam forming and MIMO to increase spectral efficiencyeven further.

When considering spectral efficiency of a practical wirelesscommunication system, due to possibly different practical bandwidthdefinitions (and also not strictly bandlimited nature of actual transmitsignal), the following approach would be more appropriate.

Referring now to FIG. 36, consider the equivalent discrete time system,and obtain the Shannon capacity for that system (will be denoted by Cd).Regarding the discrete time system, for example, for conventional QAMsystems in AWGN, the system will be:y[n]=ax[n]+w[n]where a is a scalar representing channel gain and amplitude scaling,x[n] is the input signal (QAM symbol) with unit average energy (scalingis embedded in a), y[n] is the demodulator (matched filter) outputsymbol, and index n is the discrete time index.

The corresponding Shannon capacity is:C _(d)=log₂(1+|a| ²/σ²)where σ2 is the noise variance (in complex dimension) and |a|2/σ2 is theSNR of the discrete time system.

Second, compute the bandwidth W based on the adopted bandwidthdefinition (e.g., bandwidth defined by −40 dBc out of band power). Ifthe symbol duration corresponding to a sample in discrete time (or thetime required to transmit Ca bits) is T, then the spectral efficiencycan be obtained as:C/W=Ca/(TW) bps/HzIn discrete time system in AWGN channels, using Turbo or similar codeswill give performance quite close to Shannon limit Ca. This performancein discrete time domain will be the same regardless of the pulse shapeused. For example, using either SRRC (square root raised cosine) pulseor a rectangle pulse gives the same Ca (or Ca/T). However, when weconsider continuous time practical systems, the bandwidths of SRRC andthe rectangle pulse will be different. For a typical practical bandwidthdefinition, the bandwidth for a SRRC pulse will be smaller than that forthe rectangle pulse and hence SRRC will give better spectral efficiency.In other words, in discrete time system in AWGN channels, there islittle room for improvement. However, in continuous time practicalsystems, there can be significant room for improvement in spectralefficiency.

Referring now to FIG. 37, there is illustrated a PSD plot (BLANK) ofMLO, modified MLO (MMLO) and square root raised cosine (SRRC). From theillustration in FIG. 37, demonstrates the better localization propertyof MLO. An advantage of MLO is the bandwidth. FIG. 36 also illustratesthe interferences to adjacent channels will be much smaller for MLO.This will provide additional advantages in managing, allocating orpackaging spectral resources of several channels and systems, andfurther improvement in overall spectral efficiency. If the bandwidth isdefined by the −40 dBc out of band power, the within-bandwidth PSDs ofMLO and SRRC are illustrated in FIG. 38. The ratio of the bandwidths isabout 1.536. Thus, there is significant room for improvement in spectralefficiency.

Modified MLO systems are based on block-processing wherein each blockcontains N MLO symbols and each MLO symbol has L layers. MMLO can beconverted into parallel (virtual) orthogonal channels with differentchannel SNRs as illustrated in FIG. 39. The outputs provide equivalentdiscrete time parallel orthogonal channels of MMLO.

Note that the intersymbol interference caused pulse overlapping of MLOhas been addressed by the parallel orthogonal channel conversion. As anexample, the power gain of a parallel orthogonal virtual channel of MMLOwith three layers and 40 symbols per block is illustrated in FIG. 39.FIG. 40 illustrates the channel power gain of the parallel orthogonalchannels of MMLO with three layers and T_(sim)=3. By applying a waterfilling solution, an optimal power distribution across the orthogonalchannels for a fixed transmit power may be obtained. The transmit poweron the k^(th) orthogonal channel is denoted by P_(k). Then the discretetime capacity of the MMLO can be given by:

$C_{d} = {\sum\limits_{k = 1}^{k}{{\log_{2}\left( {1 + \frac{P_{k}{a_{k}}^{2}}{\sigma_{k}^{2}}} \right)}\mspace{14mu}{bits}\mspace{14mu}{per}\mspace{14mu}{block}}}$Note that K depends on the number of MLO layers, the number of MLOsymbols per block, and MLO symbol duration.For MLO pulse duration defined by [−t₁, t₁], and symbol durationT_(mlo), the MMLO block length is:T _(block)=(N−1)T _(mlo)+2t ₁Suppose the bandwidth of MMLO signal based on the adopted bandwidthdefinition (ACLR, OBP, or other) is W_(mmlo), then the practicalspectral efficiency of MMLO is given by:

$\frac{C_{d}}{W_{mmlo}T_{block}} = {\frac{1}{W_{mmlo}\left\{ {{\left( {N - 1} \right)T_{mlo}} + {2\; t_{1}}} \right\}}{\sum\limits_{k = 1}^{K}{{\log_{2}\left( {1 + \frac{P_{k}{a_{k}}^{2}}{\sigma_{k}^{2}}} \right)}\;\frac{bps}{Hz}}}}$

FIGS. 41-42 show the spectral efficiency comparison of MMLO with N=40symbols per block, L=3 layers, T_(mlo)=3, t₁=8, and SRRC with duration[−8T, 8T], T=1, and the roll-off factor β=0.22, at SNR of 5 dB. Twobandwidth definitions based on ACLR1 (first adjacent channel leakagepower ratio) and OBP (out of band power) are used.

FIGS. 43-44 show the spectral efficiency comparison of MMLO with L=4layers. The spectral efficiencies and the gains of MMLO for specificbandwidth definitions are shown in the following tables.

TABLE D Spectral Efficiency (bps/Hz) Gain with based on ACLR1 ≤30 dBcreference to per bandwidth SRRC SRRC 1.7859 1 MMLO (3 layers, Tmlo = 3)2.7928 1.5638 MMLO (4 layers, Tmlo = 3) 3.0849 1.7274

TABLE E Gain with Spectral Efficiency (bps/Hz) reference based on OBP ≤−40 dBc to SRRC SRRC 1.7046 1 MMLO (3 layers, Tmlo = 3) 2.3030 1.3510MMLO (4 layers, Tmlo = 3) 2.6697 1.5662

Referring now to FIGS. 45 and 46, there are provided basic blockdiagrams of low-pass-equivalent MMLO transmitters (FIG. 45) andreceivers (FIG. 46). The low-pass-equivalent MMLO transmitter 4500receives a number of input signals 4502 at a block-based transmitterprocessing 4504. The transmitter processing outputs signals to theSH(L−1) blocks 4506 which produce the I&Q outputs. These signals arethen all combined together at a combining circuit 4508 for transmission.

Within the baseband receiver (FIG. 46) 4600, the received signal isseparated and applied to a series of match filters 4602. The outputs ofthe match filters are then provided to the block-based receiverprocessing block 4604 to generate the various output streams.

Consider a block of N MLO-symbols with each MLO symbol carrying Lsymbols from L layers. Then there are NL symbols in a block. Define c(m,n)=symbol transmitted by the m-th MLO layer at the n-th MLO symbol.Write all NL symbols of a block as a column vector as follows:c=[c(0,0), c(1,0), . . . , c(L−1, 0), c(0,1), c(1,1), . . . , c(L−1, 1),. . . , c(L−1, N−1)]T. Then the outputs of the receiver matched filtersfor that transmitted block in an AWGN channel, defined by the columnvector y of length NL, can be given as y=H c+n, where H is an NL×NLmatrix representing the equivalent MLO channel, and n is a correlatedGaussian noise vector.

By applying SVD to H, we have H=U D VH where D is a diagonal matrixcontaining singular values. Transmitter side processing using V and thereceiver side processing UH, provides an equivalent system with NLparallel orthogonal channels, (i.e., y=H Vc+n and UH y=Dc+UH n). Theseparallel channel gains are given by diagonal elements of D. The channelSNR of these parallel channels can be computed. Note that by thetransmit and receive block-based processing, we obtain parallelorthogonal channels and hence the ISI issue has be resolved.

Since the channel SNRs of these parallel channels are not the same, wecan apply the optimal Water filling solution to compute the transmitpower on each channel given a fixed total transmit power. Using thistransmit power and corresponding channel SNR, we can compute capacity ofthe equivalent system as given in the previous report.

Issues of Fading, Multipath, and Multi-Cell Interference

Techniques used to counteract channel fading (e.g., diversitytechniques) in conventional systems can also be applied in MMLO. Forslowly-varying multi-path dispersive channels, if the channel impulseresponse can be fed back, it can be incorporated into the equivalentsystem mentioned above, by which the channel induced ISI and theintentionally introduced MMLO ISI can be addressed jointly. For fasttime-varying channels or when channel feedback is impossible, channelequalization needs to be performed at the receiver. A block-basedfrequency-domain equalization can be applied and an oversampling wouldbe required.

If we consider the same adjacent channel power leakage for MMLO and theconventional system, then the adjacent cells' interference power wouldbe approximately the same for both systems. If interference cancellationtechniques are necessary, they can also be developed for MMLO.

Scope and System Description

This report presents the symbol error probability (or symbol error rate)performance of MLO signals in additive white Gaussian noise channel withvarious inter-symbol interference levels. As a reference, theperformance of the conventional QAM without ISI is also included. Thesame QAM size is considered for all layers of MLO and the conventionalQAM.

The MLO signals are generated from the Physicist's special Hermitefunctions:

${f_{n}\left( {t,\alpha} \right)} = {\sqrt{\frac{\alpha}{\sqrt{\pi}{n!}2^{n}}}{H_{n}\left( {\alpha\; t} \right)}e^{- \frac{\alpha^{2}t^{2}}{2}}}$where Hn(αt) is the n^(th) order Hermite polynomial. Note that thefunctions used in the lab setup correspond to

$\alpha = \frac{1}{\sqrt{2}}$and, for consistency,

$\alpha = \frac{1}{\sqrt{2}}$is used in this report.

MLO signals with 3, 4 or 10 layers corresponding to n=0-2, 0-3, or 0-9are used and the pulse duration (the range of t) is [−8, 8] in the abovefunction.

AWGN channel with perfect synchronization is considered.

The receiver consists of matched filters and conventional detectorswithout any interference cancellation, i.e., QAM slicing at the matchedfilter outputs.

${\%\mspace{14mu}{pulse}\text{-}{overlapping}} = {\frac{T_{p} - T_{sym}}{T_{p}} \times 100\%}$where Tp is the pulse duration (16 in the considered setup) and Tsym isthe reciprocal of the symbol rate in each MLO layer. The consideredcases are listed in the following table.

TABLE F % of Pulse Overlapping T_(sym) T_(p)   0% 16 16  12.5% 14 1618.75% 13 16   25% 12 16  37.5% 10 16 43.75% 9 16   50% 8 16 56.25% 7 16 62.5% 6 16   75% 4 16Derivation of the Signals Used in Modulation

To do that, it would be convenient to express signal amplitude s(t) in acomplex form close to quantum mechanical formalism. Therefore thecomplex signal can be represented as:

ψ(t) = s(t) + j σ(t) where  s(t) ≡ real  signalσ(t) = imaginary  signal  (quadrature)${\sigma(t)} = {\frac{1}{\pi}{\overset{\infty}{\int\limits_{- \infty}}{{s(\tau)}\frac{d\;\tau}{\tau - t}}}}$${s(t)} = {{- \frac{1}{\pi}}{\overset{\infty}{\int\limits_{- \infty}}{{\sigma(t)}\frac{d\;\tau}{\tau - t}}}}$Where s(t) and σ(t) are Hilbert transforms of one another and since σ(t)is qudratures of s(t), they have similar spectral components. That is ifthey were the amplitudes of sound waves, the ear could not distinguishone form from the other.

Let us also define the Fourier transform pairs as follows:

${\psi(t)} = {\frac{1}{\pi}{\overset{\infty}{\int\limits_{- \infty}}{{\varphi(f)}e^{j\;\omega\; t}{df}}}}$${\varphi(f)} = {\frac{1}{\pi}{\overset{\infty}{\int\limits_{- \infty}}{{\psi(t)}e^{{- j}\;\omega\; t}{dt}}}}$ψ^(*)(t)ψ(t) = [s(t)]² + [σ(t)]² + … ≡ signal  power

Let's also normalize all moments to M₀:

M₀ = ∫₀^(τ)s(t)dt M₀ = ∫₀^(τ)φ^(*)φ df

Then the moments are as follows:

M₀ = ∫₀^(τ)s(t)dt M₁ = ∫₀^(τ)ts(t)dt M₂ = ∫₀^(τ)t²s(t)dtM_(N − 1) = ∫₀^(τ)t^(N − 1)s(t)dt

In general, one can consider the signal s(t) be represented by apolynomial of order N, to fit closely to s(t) and use the coefficient ofthe polynomial as representation of data. This is equivalent tospecifying the polynomial in such a way that its first N “moments” M_(j)shall represent the data. That is, instead of the coefficient of thepolynomial, we can use the moments. Another method is to expand thesignal s(t) in terms of a set of N orthogonal functions φ_(k)(t),instead of powers of time. Here, we can consider the data to be thecoefficients of the orthogonal expansion. One class of such orthogonalfunctions are sine and cosine functions (like in Fourier series).

Therefore we can now represent the above moments using the orthogonalfunction ψ with the following moments:

$\overset{\_}{t} = {{\frac{\int{{\psi^{*}(t)}t{\psi(t)}{dt}}}{\int{{\psi^{*}(t)}{\psi(t)}{dt}}}\mspace{14mu}\overset{\_}{t^{2}}} = \frac{\int{{\psi^{*}(t)}t^{2}{\psi(t)}{dt}}}{\int{{\psi^{*}(t)}{\psi(t)}{dt}}}}$$\overset{\_}{t^{n}} = \frac{\int{{\psi^{*}(t)}t^{n}{\psi(t)}{dt}}}{\int{{\psi^{*}(t)}{\psi(t)}{dt}}}$Similarly,

$\overset{\_}{f} = {{\frac{\int{{\varphi^{*}(f)}f\;{\varphi(f)}{df}}}{\int{{\varphi^{*}(f)}{\varphi(f)}{df}}}\mspace{14mu}{\overset{\_}{f}}^{2}} = \frac{\int{{\varphi^{*}(f)}f^{2}{\varphi(f)}{df}}}{\int{{\varphi^{*}(f)}{\varphi(f)}{df}}}}$${\overset{\_}{f}}^{n} = \frac{\int{{\varphi^{*}(f)}f^{n}{\varphi(f)}{df}}}{\int{{\varphi^{*}(f)}{\varphi(f)}{df}}}$If we did not use complex signal, then:f=0To represent the mean values from time to frequency domains, replace:

φ(f) → ψ(t)$\left. f\rightarrow{\frac{1}{2\pi\; j}\frac{d}{dt}} \right.$These are equivalent to somewhat mysterious rule in quantum mechanicswhere classical momentum becomes an operator:

$\left. P_{x}\rightarrow\frac{h}{2\pi\; j} \right.\frac{\partial}{\partial x}$Therefore using the above substitutions, we have:

$\overset{¯}{f} = {\frac{\int{{\varphi^{*}(f)}f\;{\varphi(f)}df}}{\int{{\varphi^{*}(f)}{\varphi(f)}df}} = {\frac{\int{{\psi^{*}(t)}\left( \frac{1}{2\pi j} \right)\frac{d{\psi(t)}}{dt}dt}}{\int{{\psi^{*}(t)}{\psi(t)}dt}} = {\left( \frac{1}{2\pi j} \right)\frac{\int{\psi^{*}\frac{d\psi}{dt}dt}}{\int{\psi^{*}\psi dt}}}}}$  And:${\overset{\_}{f}}^{2} = {\frac{\int{{\varphi^{*}(f)}f^{2}{\varphi(f)}df}}{\int{{\varphi^{*}(f)}{\varphi(f)}df}} = {\frac{\int{{\psi^{*}\left( \frac{1}{2\pi j} \right)}^{2}\frac{d^{2}}{dt^{2}}\psi dt}}{\int{\psi^{*}\psi dt}} = {{- \left( \frac{1}{2\pi} \right)^{2}}\frac{\int{\psi^{*}\frac{d^{2}}{dt^{2}}\psi dt}}{\int{\psi^{*}\psi dt}}}}}$$\mspace{20mu}{\overset{\_}{t^{2}} = \frac{\int{\psi^{*}t^{2}{\psi dt}}}{\int{\psi^{*}{\psi dt}}}}$We can now define an effective duration and effective bandwidth as:

${\Delta t} = {\sqrt{2\pi\;\overset{\_}{\left( {t - \overset{\_}{t}} \right)^{2}}} = {2{\pi \cdot {rms}}\mspace{14mu}{in}\mspace{14mu}{time}}}$${\Delta\; f} = {\sqrt{2\pi\;\overset{\_}{\left( {f - \overset{¯}{f}} \right)^{2}}} = {2{\pi \cdot {rms}}\mspace{14mu}{in}\mspace{14mu}{frequency}}}$But we know that:(t−t)² = t ² −( t )²(f−f)² = f ² −( f )²We can simplify if we make the following substitutions:τ=t−tΨ(τ)=ψ(t)e ^(−jωτ)ω₀=ω=2π f=2πf ₀We also know that:(Δt)²(Δf)²=(ΔtΔf)²And therefore:

$\left( {\Delta t\Delta f} \right)^{2} = {{\frac{1}{4}\left\lbrack {4\frac{\int{{\Psi^{*}(\tau)}\tau^{2}{\Psi(\tau)}d\tau{\int{\frac{d\Psi^{*}}{d\;\tau}\frac{d\;\Psi}{d\;\tau}d\;\tau}}}}{\left( {\int{{\Psi^{*}(\tau)}{\psi(\tau)}d\tau}} \right)^{2}}} \right\rbrack} \geq \left( \frac{1}{4} \right)}$$\left( {\Delta t\Delta f} \right) \geq \left( \frac{1}{2} \right)$Now instead of

$\left( {\Delta t\Delta f} \right) \geq \left( \frac{1}{2} \right)$we are interested to force the equality

$\left( {\Delta t\Delta f} \right) = \left( \frac{1}{2} \right)$and see what signals satisfy the equality. Given the fixed bandwidth Δf,the most efficient transmission is one that minimizes the time-bandwidthproduct

$\left( {\Delta t\Delta f} \right) = \left( \frac{1}{2} \right)$For a given bandwidth Δf, the signal that minimizes the transmission inminimum time will be a Gaussian envelope. However, we are often givennot the effective bandwidth, but always the total bandwidth f₂−f₁. Now,what is the signal shape which can be transmitted through this channelin the shortest effective time and what is the effective duration?

$\left. {{\Delta t}==\frac{\frac{1}{\left( {2\pi} \right)^{2}}{\int_{f_{1}}^{f_{2}}{\frac{d\;\varphi^{*}}{df}\frac{d\;\varphi}{df}}}}{\int_{f_{1}}^{f_{2}}{\varphi^{*}\varphi\;{df}}}}\rightarrow\min \right.$Where φ(f) is zero outside the range f₂−f₁.

To do the minimization, we would use the calculus of variations(Lagrange's Multiplier technique). Note that the denominator is constantand therefore we only need to minimize the numerator as:

$\mspace{20mu}{\left. {\Delta\; t}\rightarrow\left. \min\rightarrow{\delta{\int_{f_{1}}^{f_{2}}{\left( {{\frac{d\;\varphi^{*}}{df}\frac{d\;\varphi}{df}} + {{\Lambda\varphi}^{*}\varphi}} \right){df}}}} \right. \right. = 0}$  First  Trem${\delta{\int_{f_{1}}^{f_{2}}{\frac{d\;\varphi^{*}}{df}\frac{d\;\varphi}{df}df}}} = {{\int{\left( {{\frac{d\;\varphi^{*}}{df}\delta\frac{d\;\varphi}{df}} + {\frac{d\;\varphi}{df}\delta\frac{d\;\varphi^{*}}{df}}} \right)df}} = {{\int{\left( {{\frac{d\;\varphi^{*}}{df}\frac{d\;\delta\;\varphi}{df}} + {\frac{d\;\varphi}{df}\frac{d\;{\delta\varphi}^{*}}{df}}} \right){df}}} = {{\left\lbrack {{\frac{d\;\varphi^{*}}{df}{\delta\varphi}} + {\frac{d\;\varphi}{df}{\delta\varphi}^{*}}} \right\rbrack_{f_{1}}^{f_{2}} - {\int{\left( {{\frac{d^{2}\varphi^{*}}{df^{2}}{\delta\varphi}} + {\frac{d^{2}\varphi}{df^{2}}{\delta\varphi}^{*}}} \right){df}}}} = {\int{\left( {{\frac{d^{2}\varphi^{*}}{df^{2}}{\delta\varphi}} + {\frac{d^{2}\varphi}{df^{2}}{\delta\varphi}^{*}}} \right){df}}}}}}$  Second  Trem  δ∫_(f₁)^(f₂)(Λφ^(*)φ)df = Λ∫_(f₁)^(f₂)(φ^(*)δφ + φδφ^(*))df$\mspace{20mu}{{BothTrems} = {{\int{\left\lbrack {{\left( {\frac{d^{2}\varphi^{*}}{df^{2}} + {\Lambda\varphi}^{*}} \right){\delta\varphi}} + {\left( {\frac{d^{2}\varphi}{df^{2}} + {\Lambda\varphi}} \right){\delta\varphi}^{*}}} \right\rbrack{df}}} = 0}}$This is only possible if and only if:

$\left( {\frac{d^{2}\varphi}{df^{2}} + {\Lambda\varphi}} \right) = 0$The solution to this is of the form

${\varphi(f)} = {\sin\; k\;{\pi\left( \frac{f - f_{1}}{f_{2} - f_{1}} \right)}}$Now if we require that the wave vanishes at infinity, but still satisfythe minimum time-bandwidth product:

$\left( {\Delta t\Delta f} \right) = \left( \frac{1}{2} \right)$Then we have the wave equation of a Harmonic Oscillator:

${\frac{d^{2}{\Psi(\tau)}}{d\tau^{2}} + {\left( {\lambda - {\alpha^{2}\tau^{2}}} \right){\Psi(\tau)}}} = 0$which vanishes at infinity only if:

λ = α(2n + 1)$\psi_{n} = {{e^{{- \frac{1}{2}}\omega^{2}\tau^{2}}\frac{d^{n}}{d\;\tau^{n}}e^{{- \alpha^{2}}\tau^{2}}} \propto {H_{n}(\tau)}}$Where H_(n)(τ) is the Hermit functions and:(ΔtΔf)=½(2n+1)So Hermit functions H_(n)(τ) occupy information blocks of 1/2, 3/2, 5/2,. . . with 1/2 as the minimum information quanta.Squeezed States

Here we would derive the complete Eigen functions in the mostgeneralized form using quantum mechanical approach of Dirac algebra. Westart by defining the following operators:

${{b = {\sqrt{\frac{m\;\omega^{\prime}}{2\hslash}}\left( {x + \frac{ip}{m\;\omega^{\prime}}} \right)}}b^{+}} = {{\sqrt{\frac{m\;\omega^{\prime}}{2\hslash}}{\left( {x - \frac{ip}{m\;\omega^{\prime}}} \right)\left\lbrack {b,b^{+}} \right\rbrack}} = 1}$a = λb − μb⁺ a⁺ = λb⁺ − μbNow we are ready to define Δx and Δp as:

${\left( {\Delta x} \right)^{2} = {{\frac{\hslash}{2m\;\omega}\left( \frac{\omega}{\omega^{\prime}} \right)} = {\frac{\hslash}{2m\;\omega}\left( {\lambda - \mu} \right)^{2}}}}{\left( {\Delta p} \right)^{2} = {{\frac{\hslash\; m\;\omega}{2}\left( \frac{\omega^{\prime}}{\omega} \right)} = {\frac{\hslash\; m\;\omega}{2}\left( {\lambda + \mu} \right)^{2}}}}{{\left( {\Delta x} \right)^{2}\left( {\Delta p} \right)^{2}} = {\frac{\hslash^{2}}{4}\left( {\lambda^{2} - \mu^{2}} \right)^{2}}}{{\Delta x\Delta p} = {{\frac{\hslash}{2}\left( {\lambda^{2} - \mu^{2}} \right)} = \frac{\hslash}{2}}}$Now let parameterize differently and instead of two variables λ and μ,we would use only one variable ξ as follows:λ=sin hξμ=cos hξλ+μ=e ^(ξ)λ−μ=−e ^(−ξ)Now the Eigen states of the squeezed case are:

bβ⟩ = ββ⟩ (λa + μa⁺)β⟩ = ββ⟩ b = UaU⁺ U = e^(ξ/2(a²a^(+²)))U⁺(ξ)aU(ξ) = acosh ξ − a⁺sinh ξ U⁺(ξ)a⁺U(ξ) = a⁺cosh ξ − asinh ξWe can now consider the squeezed operator:

α, ξ⟩ = U(ξ)D(α)0⟩${D(\alpha)} = {e^{\frac{- {|\alpha|^{2}}}{2}}e^{\alpha\; a^{+}}e^{{- {\alpha\;}^{*}}a}}$$\left. \alpha \right\rangle = {\sum\limits_{n = 0}^{\infty}{\frac{\alpha^{n}}{\sqrt{n!}}e\left. n \right\rangle}}$$\left. \alpha \right\rangle = {e^{\frac{- {\alpha }^{2}}{2} + {\alpha\; a^{+}}}\left. 0 \right\rangle}$For a distribution P(n) we would have:

P(n) = ⟨nβ, ξ⟩²$\left\langle {{\alpha{}\beta},\xi} \right\rangle = {\sum\limits_{n = 0}^{\infty}{\frac{\alpha^{n}}{\sqrt{n!}}e^{\frac{- {\alpha }^{2}}{2}}\left\langle {{n{}\beta},\xi} \right\rangle}}$$e^{{2{zt}} - t^{2}} = {\sum\limits_{n = 0}^{\infty}\frac{{H_{n}(z)}t^{n}}{n!}}$Therefore the final result is:

$\begin{matrix}{\left\langle {{n{}\beta},\xi} \right\rangle = {\frac{\left( {\tanh\xi} \right)^{n/2}}{2^{n/2}\left( {n{!{\cosh\xi}}} \right)^{2}}e^{{{- 1}/2}{({{\beta }^{2} - {\beta^{2}\tanh\;\xi}})}}{H_{n}\left( \frac{\beta}{2\sinh\xi\cosh\xi} \right)}}}\end{matrix}$Free Space Communications

An additional configuration in which the optical angular momentumprocessing and multi-layer overlay modulation technique described hereinabove may prove useful within the optical network framework is use withfree-space optics communications. Free-space optics systems provide anumber of advantages over traditional UHF RF based systems from improvedisolation between the systems, the size and the cost of thereceivers/transmitters, lack of RF licensing laws, and by combiningspace, lighting, and communication into the same system. Referring nowto FIG. 47, there is illustrated an example of the operation of afree-space communication system. The free-space communication systemutilizes a free-space optics transmitter 4702 that transmits alight beam4706 to a free-space optics receiver 4704. The major difference betweena fiber-optic network and a free-space optic network is that theinformation beam is transmitted through free space rather than over afiber-optic cable. This causes a number of link difficulties, which willbe more fully discussed herein below. Free-space optics is a line ofsight technology that uses the invisible beams of light to provideoptical bandwidth connections that can send and receive up to 2.5 Gbpsof data, voice, and video communications between a transmitter 4702 anda receiver 4704. Free-space optics uses the same concepts asfiber-optics, except without the use of a fiber-optic cable. Free-spaceoptics systems provide the light beam 4706 within the infrared (IR)spectrum, which is at the low end of the light spectrum. Specifically,the optical signal is in the range of 300 Gigahertz to 1 Terahertz interms of wavelength.

Presently existing free-space optics systems can provide data rates ofup to 10 Gigabits per second at a distance of up to 2.5 kilometers. Inouter space, the communications range of free space opticalcommunications is currently on the order of several thousand kilometers,but has the potential to bridge interplanetary distances of millions ofkilometers, using optical telescopes as beam expanders. In January of2013, NASA used lasers to beam an image of the Mona Lisa to the LunarReconnaissance Orbiter roughly 240,000 miles away. To compensate foratmospheric interference, an error correction code algorithm, similar tothat used within compact discs, was implemented.

The distance records for optical communications involve detection andemission of laser light by space probes. A two-way distance record forcommunication was established by the Mercury Laser Altimeter instrumentaboard the MESSENGER spacecraft. This infrared diode neodymium laser,designed as a laser altimeter for a Mercury Orbiter mission, was able tocommunicate across a distance of roughly 15,000,000 miles (24,000,000kilometers) as the craft neared Earth on a fly by in May of 2005. Theprevious record had been set with a one-way detection of laser lightfrom Earth by the Galileo Probe as two ground based lasers were seenfrom 6,000,000 kilometers by the outbound probe in 1992. Researchersused a white LED based space lighting system for indoor local areanetwork communications.

Referring now to FIG. 48, there is illustrated a block diagram of afree-space optics system using orbital angular momentum and multileveloverlay modulation according to the present disclosure. While thepresent disclosure is made with respect to a system using OAM and MLOmodulation, it will be realized that a system can implement only one ofor neither of these techniques. The OAM twisted signals, in addition tobeing transmitted over fiber, may also be transmitted using free optics.In this case, the transmission signals are generated within transmissioncircuitry 4802 at each of the FSO transceivers 4804. Free-space opticstechnology is based on the connectivity between the FSO based opticalwireless units, each consisting of an optical transceiver 4804 with atransmitter 4802 and a receiver 4806 to provide full duplex open pairand bidirectional closed pairing capability. Each optical wirelesstransceiver unit 4804 additionally includes an optical source 4808 plusa lens or telescope 4810 for transmitting light through the atmosphereto another lens 4810 receiving the information. At this point, thereceiving lens or telescope 4810 connects to a high sensitivity receiver4806 via optical fiber 4812. The transmitting transceiver 4804 a and thereceiving transceiver 4804 b have to have line of sight to each other.Trees, buildings, animals, and atmospheric conditions all can hinder theline of sight needed for this communications medium. Since line of sightis so critical, some systems make use of beam divergence or a diffusedbeam approach, which involves a large field of view that toleratessubstantial line of sight interference without significant impact onoverall signal quality. The system may also be equipped with autotracking mechanism 4814 that maintains a tightly focused beam on thereceiving transceiver 3404 b, even when the transceivers are mounted ontall buildings or other structures that sway.

The modulated light source used with optical source 4808 is typically alaser or light emitting diode (LED) providing the transmitted opticalsignal that determines all the transmitter capabilities of the system.Only the detector sensitivity within the receiver 4806 plays an equallyimportant role in total system performance. For telecommunicationspurposes, only lasers that are capable of being modulated at 20 Megabitsper second to 2.5 Gigabits per second can meet current marketplacedemands. Additionally, how the device is modulated and how muchmodulated power is produced are both important to the selection of thedevice. Lasers in the 780-850 nm and 1520-1600 nm spectral bands meetfrequency requirements.

Commercially available FSO systems operate in the near IR wavelengthrange between 750 and 1600 nm, with one or two systems being developedto operate at the IR wavelength of 10,000 nm. The physics andtransmissions properties of optical energy as it travels through theatmosphere are similar throughout the visible and near IR wavelengthrange, but several factors that influence which wavelengths are chosenfor a particular system.

The atmosphere is considered to be highly transparent in the visible andnear IR wavelength. However, certain wavelengths or wavelength bands canexperience severe absorption. In the near IR wavelength, absorptionoccurs primarily in response to water particles (i.e., moisture) whichare an inherent part of the atmosphere, even under clear weatherconditions. There are several transmission windows that are nearlytransparent (i.e., have an attenuation of less than 0.2 dB perkilometer) within the 700-10,000 nm wavelength range. These wavelengthsare located around specific center wavelengths, with the majority offree-space optics systems designed to operate in the windows of 780-850nm and 1520-1600 nm.

Wavelengths in the 780-850 nm range are suitable for free-space opticsoperation and higher power laser sources may operate in this range. At780 nm, inexpensive CD lasers may be used, but the average lifespan ofthese lasers can be an issue. These issues may be addressed by runningthe lasers at a fraction of their maximum rated output power which willgreatly increase their lifespan. At around 850 nm, the optical source4808 may comprise an inexpensive, high performance transmitter anddetector components that are readily available and commonly used innetwork transmission equipment. Highly sensitive silicon (SI) avalanchephotodiodes (APD) detector technology and advanced vertical cavityemitting laser may be utilized within the optical source 4808.

VCSEL technology may be used for operation in the 780 to 850 nm range.Possible disadvantage of this technology include beam detection throughthe use of a night vision scope, although it is still not possible todemodulate a perceived light beam using this technique.

Wavelengths in the 1520-1600 nm range are well-suited for free-spacetransmission, and high quality transmitter and detector components arereadily available for use within the optical source block 4808. Thecombination of low attenuation and high component availability withinthis wavelength range makes the development of wavelength divisionmultiplexing (WDM) free-space optics systems feasible. However,components are generally more expensive and detectors are typically lesssensitive and have a smaller receive surface area when compared withsilicon avalanche photodiode detectors that operator at the 850 nmwavelength. These wavelengths are compatible with erbium-doped fiberamplifier technology, which is important for high power (greater than500 milliwatt) and high data rate (greater than 2.5 Gigabytes persecond) systems. Fifty to 65 times as much power can be transmitted atthe 1520-1600 nm wavelength than can be transmitted at the 780-850 nmwavelength for the same eye safety classification. Disadvantages ofthese wavelengths include the inability to detect a beam with a nightvision scope. The night vision scope is one technique that may be usedfor aligning the beam through the alignment circuitry 4814. Class 1lasers are safe under reasonably foreseeable operating conditionsincluding the use of optical instruments for intrabeam viewing. Class 1systems can be installed at any location without restriction.

Another potential optical source 4808 comprised Class 1M lasers. Class1M laser systems operate in the wavelength range from 302.5 to 4000 nm,which is safe under reasonably foreseeable conditions, but may behazardous if the user employs optical instruments within some portion ofthe beam path. As a result, Class 1M systems should only be installed inlocations where the unsafe use of optical aids can be prevented.Examples of various characteristics of both Class 1 and Class 1M lasersthat may be used for the optical source 4808 are illustrated in Table Gbelow.

TABLE G Laser Power Aperture Size Distance Power Density Classification(mW) (mm) (m) (mW/cm²) 850-nm Wavelength Class 1 0.78 7 14 2.03 50 20000.04 Class 1M 0.78 7 100 2.03 500 7 14 1299.88 50 2000 25.48 1550-nmWavelength Class 1 10 7 14 26.00 25 2000 2.04 Class 1M 10 3.5 100 103.99500 7 14 1299.88 25 2000 101.91

The 10,000 nm wavelength is relatively new to the commercial free spaceoptic arena and is being developed because of better fog transmissioncapabilities. There is presently considerable debate regarding thesecharacteristics because they are heavily dependent upon fog type andduration. Few components are available at the 10,000 nm wavelength, asit is normally not used within telecommunications equipment.Additionally, 10,000 nm energy does not penetrate glass, so it isill-suited to behind window deployment.

Within these wavelength windows, FSO systems should have the followingcharacteristics. The system should have the ability to operate at higherpower levels, which is important for longer distance FSO systemtransmissions. The system should have the ability to provide high speedmodulation, which is important for high speed FSO systems. The systemshould provide a small footprint and low power consumption, which isimportant for overall system design and maintenance. The system shouldhave the ability to operate over a wide temperature range without majorperformance degradations such that the systems may prove useful foroutdoor systems. Additionally, the mean time between failures shouldexceed 10 years. Presently existing FSO systems generally use VCSELS foroperation in the shorter IR wavelength range, and Fabry-Perot ordistributed feedback lasers for operation in the longer IR wavelengthrange. Several other laser types are suitable for high performance FSOsystems.

A free-space optics system using orbital angular momentum processing andmulti-layer overlay modulation would provide a number of advantages. Thesystem would be very convenient. Free-space optics provides a wirelesssolution to a last-mile connection, or a connection between twobuildings. There is no necessity to dig or bury fiber cable. Free-spaceoptics also requires no RF license. The system is upgradable and itsopen interfaces support equipment from a variety of vendors. The systemcan be deployed behind windows, eliminating the need for costly rooftopright. It is also immune to radiofrequency interference or saturation.The system is also fairly speedy. The system provides 2.5 Gigabits persecond of data throughput. This provides ample bandwidth to transferfiles between two sites. With the growth in the size of files,free-space optics provides the necessary bandwidth to transfer thesefiles efficiently.

Free-space optics also provides a secure wireless solution. The laserbeam cannot be detected with a spectral analyzer or RF meter. The beamis invisible, which makes it difficult to find. The laser beam that isused to transmit and receive the data is very narrow. This means that itis almost impossible to intercept the data being transmitted. One wouldhave to be within the line of sight between the receiver and thetransmitter in order to be able to accomplish this feat. If this occurs,this would alert the receiving site that a connection has been lost.Thus, minimal security upgrades would be required for a free-spaceoptics system.

However, there are several weaknesses with free-space optics systems.The distance of a free-space optics system is very limited. Currentlyoperating distances are approximately within 2 kilometers. Although thisis a powerful system with great throughput, the limitation of distanceis a big deterrent for full-scale implementation. Additionally, allsystems require line of sight be maintained at all times duringtransmission. Any obstacle, be it environmental or animals can hinderthe transmission. Free-space optic technology must be designed to combatchanges in the atmosphere which can affect free-space optic systemperformance capacity.

Something that may affect a free-space optics system is fog. Dense fogis a primary challenge to the operation of free-space optics systems.Rain and snow have little effect on free-space optics technology, butfog is different. Fog is a vapor composed of water droplets which areonly a few hundred microns in diameter, but can modify lightcharacteristics or completely hinder the passage of light through acombination of absorption, scattering, and reflection. The primaryanswer to counter fog when deploying free-space optic based wirelessproducts is through a network design that shortens FSO linked distancesand adds network redundancies.

Absorption is another problem. Absorption occurs when suspended watermolecules in the terrestrial atmosphere extinguish photons. This causesa decrease in the power density (attenuation) of the free space opticsbeam and directly affects the availability of the system. Absorptionoccurs more readily at some wavelengths than others. However, the use ofappropriate power based on atmospheric conditions and the use of spatialdiversity (multiple beams within an FSO based unit), helps maintain therequired level of network availability.

Solar interference is also a problem. Free-space optics systems use ahigh sensitivity receiver in combination with a larger aperture lens. Asa result, natural background light can potentially interfere withfree-space optics signal reception. This is especially the case with thehigh levels of background radiation associated with intense sunlight. Insome instances, direct sunlight may case link outages for periods ofseveral minutes when the sun is within the receiver's field of vision.However, the times when the receiver is most susceptible to the effectsof direct solar illumination can be easily predicted. When directexposure of the equipment cannot be avoided, the narrowing of receiverfield of vision and/or using narrow bandwidth light filters can improvesystem performance. Interference caused by sunlight reflecting off of aglass surface is also possible.

Scattering issues may also affect connection availability. Scattering iscaused when the wavelength collides with the scatterer. The physicalsize of the scatterer determines the type of scattering. When thescatterer is smaller than the wavelength, this is known as Rayleighscattering. When a scatterer is of comparable size to the wavelengths,this is known as Mie scattering. When the scattering is much larger thanthe wavelength, this is known as non-selective scattering. Inscattering, unlike absorption, there is no loss of energy, only adirectional redistribution of energy that may have significant reductionin beam intensity over longer distances.

Physical obstructions such as flying birds or construction cranes canalso temporarily block a single beam free space optics system, but thistends to cause only short interruptions. Transmissions are easily andautomatically resumed when the obstacle moves. Optical wireless productsuse multibeams (spatial diversity) to address temporary abstractions aswell as other atmospheric conditions, to provide for greateravailability.

The movement of buildings can upset receiver and transmitter alignment.Free-space optics based optical wireless offerings use divergent beamsto maintain connectivity. When combined with tracking mechanisms,multiple beam FSO based systems provide even greater performance andenhanced installation simplicity.

Scintillation is caused by heated air rising from the Earth or man-madedevices such as heating ducts that create temperature variations amongdifferent pockets of air. This can cause fluctuations in signalamplitude, which leads to “image dancing” at the free-space optics basedreceiver end. The effects of this scintillation are called “refractiveturbulence.” This causes primarily two effects on the optical beams.Beam wander is caused by the turbulent eddies that are no larger thanthe beam. Beam spreading is the spread of an optical beam as itpropagates through the atmosphere.

Referring now to FIGS. 49A through 49D, in order to achieve higher datacapacity within optical links, an additional degree of freedom frommultiplexing multiple data channels must be exploited. Moreover, theability to use two different orthogonal multiplexing techniques togetherhas the potential to dramatically enhance system performance andincreased bandwidth.

One multiplexing technique which may exploit the possibilities is modedivision multiplexing (MDM) using orbital angular momentum (OAM). OAMmode refers to laser beams within a free-space optical system orfiber-optic system that have a phase term of e^(ilφ) in their wavefronts, in which φ is the azimuth angle and l determines the OAM value(topological charge). In general, OAM modes have a “donut-like” ringshaped intensity distribution. Multiple spatial collocated laser beams,which carry different OAM values, are orthogonal to each other and canbe used to transmit multiple independent data channels on the samewavelength. Consequently, the system capacity and spectral efficiency interms of bits/S/Hz can be dramatically increased. Free-spacecommunications links using OAM may support 100 Tbits/capacity. Varioustechniques for implementing this as illustrated in FIGS. 49A through 49Dinclude a combination of multiple beams 4902 having multiple differentOAM values 4904 on each wavelength. Thus, beam 4902 includes OAM values,OAM1 and OAM4. Beam 4906 includes OAM value 2 and OAM value 5. Finally,beam 4908 includes OAM3 value and OAM6 value. Referring now to FIG. 48B,there is illustrated a single beam wavelength 4910 using a first groupof OAM values 4912 having both a positive OAM value 4912 and a negativeOAM value 4914. Similarly, OAM2 value may have a positive value 4916 anda negative value 4918 on the same wavelength 4910.

FIG. 49C illustrates the use of a wavelength 4920 having polarizationmultiplexing of OAM value. The wavelength 4920 can have multiple OAMvalues 4922 multiplexed thereon. The number of available channels can befurther increased by applying left or right handed polarization to theOAM values. Finally, FIG. 49D illustrates two groups of concentric rings4960, 4962 for a wavelength having multiple OAM values.

Wavelength distribution multiplexing (WDM) has been widely used toimprove the optical communication capacity within both fiber-opticsystems and free-space communication system. OAM mode multiplexing andWDM are mutually orthogonal such that they can be combined to achieve adramatic increase in system capacity. Referring now to FIG. 50, there isillustrated a scenario where each WDM channel 5002 contains manyorthogonal OAM beam 5004. Thus, using a combination of orbital angularmomentum with wave division multiplexing, a significant enhancement incommunication link to capacity may be achieved.

Current optical communication architectures have considerable routingchallenges. A routing protocol for use with free-space optic system musttake into account the line of sight requirements for opticalcommunications within a free-space optics system. Thus, a free-spaceoptics network must be modeled as a directed hierarchical random sectorgeometric graph in which sensors route their data via multi-hop paths toa base station through a cluster head. This is a new efficient routingalgorithm for local neighborhood discovery and a base station uplink anddownlink discovery algorithm. The routing protocol requires order Olog(n) storage at each node versus order O(n) used within currenttechniques and architectures.

Current routing protocols are based on link state, distance vectors,path vectors, or source routing, and they differ from the new routingtechnique in significant manners. First, current techniques assume thata fraction of the links are bidirectional. This is not true within afree-space optic network in which all links are unidirectional. Second,many current protocols are designed for ad hoc networks in which therouting protocol is designed to support multi-hop communications betweenany pair of nodes. The goal of the sensor network is to route sensorreadings to the base station. Therefore, the dominant traffic patternsare different from those in an ad hoc network. In a sensor network, nodeto base stations, base station to nodes, and local neighborhoodcommunication are mostly used.

Recent studies have considered the effect of unidirectional links andreport that as many as 5 percent to 10 percent of links and wireless adhoc networks are unidirectional due to various factors. Routingprotocols such as DSDV and AODV use a reverse path technique, implicitlyignoring such unidirectional links and are therefore not relevant inthis scenario. Other protocols such as DSR, ZRP, or ZRL have beendesigned or modified to accommodate unidirectionality by detectingunidirectional links and then providing bidirectional abstraction forsuch links. Referring now to FIG. 51, the simplest and most efficientsolution for dealing with unidirectionality is tunneling, in whichbidirectionality is emulated for a unidirectional link by usingbidirectional links on a reverse back channel to establish the tunnel.Tunneling also prevents implosion of acknowledgement packets and loopingby simply pressing link layer acknowledgements for tunneled packetsreceived on a unidirectional link. Tunneling, however, works well inmostly bidirectional networks with few unidirectional links.

Within a network using only unidirectional links such as a free-spaceoptical network, systems such as that illustrated in FIGS. 51 and 52would be more applicable. Nodes within a unidirectional network utilizea directional transmit 5102 transmitting from the node 5100 in a single,defined direction. Additionally, each node 5100 includes anomnidirectional receiver 5004 which can receive a signal coming to thenode in any direction. Also, as discussed here and above, the node 5000would also include a 0 log(n) storage 5106. Thus, each node 5100 provideonly unidirectional communications links. Thus, a series of nodes 5200as illustrated in FIG. 52 may unidirectionally communicate with anyother node 5200 and forward communication from one desk location toanother through a sequence of interconnected nodes.

Topological charge may be multiplexed to the wave length for eitherlinear or circular polarization. In the case of linear polarizations,topological charge would be multiplexed on vertical and horizontalpolarization. In case of circular polarization, topological charge wouldbe multiplexed on left hand and right hand circular polarizations.

The topological charges can be created using Spiral Phase Plates (SPPs)such as that illustrated in FIG. 17E, phase mask holograms or a SpatialLight Modulator (SLM) by adjusting the voltages on SLM which createsproperly varying index of refraction resulting in twisting of the beamwith a specific topological charge. Different topological charges can becreated and muxed together and de-muxed to separate charges.

As Spiral Phase plates can transform a plane wave (l=0) to a twistedwave of a specific helicity (i.e. l=+1), Quarter Wave Plates (QWP) cantransform a linear polarization (s=0) to circular polarization (i.e.s=+1).

Cross talk and multipath interference can be reduced usingMultiple-Input-Multiple-Output (MIMO).

Most of the channel impairments can be detected using a control or pilotchannel and be corrected using algorithmic techniques (closed loopcontrol system).

Multiplexing of the topological charge to the RF as well as free spaceoptics in real time provides redundancy and better capacity. Whenchannel impairments from atmospheric disturbances or scintillationimpact the information signals, it is possible to toggle between freespace optics to RF and back in real time. This approach still usestwisted waves on both the free space optics as well as the RF signal.Most of the channel impairments can be detected using a control or pilotchannel and be corrected using algorithmic techniques (closed loopcontrol system) or by toggling between the RF and free space optics.

In a further embodiment illustrated in FIG. 53, both RF signals and freespace optics may be implemented within a dual RF and free space opticsmechanism 5302. The dual RF and free space optics mechanism 5302 includea free space optics projection portion 5304 that transmits a light wavehaving an orbital angular momentum applied thereto with multileveloverlay modulation and a RF portion 5306 including circuitry necessaryfor transmitting information with orbital angular momentum andmultilayer overlay on an RF signal 5310. The dual RF and free spaceoptics mechanism 5302 may be multiplexed in real time between the freespace optics signal 5308 and the RF signal 5310 depending upon operatingconditions. In some situations, the free space optics signal 5308 wouldbe most appropriate for transmitting the data. In other situations, thefree space optics signal 5308 would not be available and the RF signal5310 would be most appropriate for transmitting data. The dual RF andfree space optics mechanism 5302 may multiplex in real time betweenthese two signals based upon the available operating conditions.

Multiplexing of the topological charge to the RF as well as free spaceoptics in real time provides redundancy and better capacity. Whenchannel impairments from atmospheric disturbances or scintillationimpact the information signals, it is possible to toggle between freespace optics to RF and back in real time. This approach still usestwisted waves on both the free space optics as well as the RF signal.Most of the channel impairments can be detected using a control or pilotchannel and be corrected using algorithmic techniques (closed loopcontrol system) or by toggling between the RF and free space optics.

Referring now to FIG. 54, there is illustrated an alternative embodimentwherein rather than using a VCSEL for transmission of the signal througha window or wall, a horn or conical antenna is used for the transmissionof signals through the window or wall. The signals transmitted via thehorn antennas are amplified for transmission in order to overcome thelosses caused by transmission of the signals through the window/wall.The device provides an optical or RF tunnel through the window or wallwithout requiring the drilling of any holes. The millimeter wavetransmission system 5402 includes an exterior portion 5404 located on anexterior of a window or wall 5406 and an interior portion 5408 locatedon the interior of the wall or window. The exterior portion 5404includes an antenna 5410 for transmitting and receiving signals to anexterior source. In a preferred embodiment, the antenna comprises a 28GHz antenna. However, it will be realized by one skilled in the art thatother antenna operating bandwidths may be utilized.

The transmitted and received signals are processed at a 28 GHzcirculator 5412. The circulator 5412 comprises an RF switch forswitching between three ports within the exterior portion 5404 and hasgood isolation. Within the circulator 5412 signals input at port 2 areoutput at port 3 and signals input at port 1 are output to port 2. Thus,the signals received by the antenna 5410 are provided to port 2 of thecirculator 5412 and output to port 3. The port 3 signals are provided tothe input of a power amplifier 5414. Similarly, the output of a poweramplifier 5416 is connected to input port 1 such that signals to betransmitted are provided to port 2 of the circulator 5412 fortransmission by antenna 5410.

The power amplifier 5412 boosts the signal strength for transmissionthrough the window or wall. The signals output from the power amplifier5414 are provided to a horn antenna 5418. The horn antenna 5418transmits to the RF signals provided from the power amplifier 5414through the window or wall 5406 to a receiving horn antenna 5420. Thehorn antennas may transmit/receive over a wide frequency band from 24GHz up to e-band. Within this range a particular band of operation forthe horn antennas is utilized. These bands include but are not limitedto 24 GHz band; 28 GHz A1 band; 28 GHz B1, A3 and B2 bands; 31 GHz bandand 39 GHz band. The horn antennas may also be of different sizes toprovide for example 10 db or 20 dB of gain.

The received signals are output from the horn antenna 5420 todemodulator circuit 5422 for demodulation. The demodulator 5422, inaddition to receiving the receive signal from for an antenna 5420,receives a signal output from a phase locked loop/local oscillator 5424.The phase locked loop/local oscillator 5424 is controlled responsive toa clock generation circuit 5426. The demodulated signal is provided fromthe demodulator 5422 to analog-to-digital converter 5428 to generate adigital output. The digital signal is routed via a router 5432 to theappropriate receiving party within the structure.

Signals to be transmitted are received from inside the building at therouter 5430. The router 5430 provides digital signals to a digital toanalog converter 5432 that converts the digital data signals into ananalog format. The analog signals are next modulated by a modulator5434. The modulator 5434 modulates the signals responsive to input fromthe phase locked loop/local oscillator 5424 under control of the clockgeneration circuit 5426. The modulated signals from modulator 5434 aretransmitted through the window/wall 5406 using a horn antenna 5436. Thesignals transmitted by horn antenna 5436 are received by a receivinghorn antenna 5438 located on the outside. The output of the horn antenna5438 is provided to the input of power amplifier 5416 that amplifies thesignal for transmission from the antenna 5410 after passing throughcirculator 5412. While the above discussion has been made with respectto the use of horn antennas for transmission through the window/wall,conical antennas may also be used for the transmissions through thewindow or wall.

Referring now to FIG. 55, there is illustrated the downlink lossesbetween the transmitting antenna 5410 and the receiving circuitry withinthe inside portion 5408. The signal is received at −110 dBm. Thereceiving antenna has a gain of 45 dB and a loss of 2 dB. Thus, thesignal output from the receiving antenna 5410 has a strength of −67 dBm.The circulator 5412 has a 2 dB loss, and the signal from the circulator5412 has a strength of −69 dBm. The power amplifier 5414 provides a 27dB to boost the signal to −42 dBm for transmission across thewindow/wall. The horn antenna 5418 provides a gain of 10 dBi to transmitthe signal at 32 dBm. The window/wall provides an approximately 40 dBloss. The receive horn antenna 5420 receives the signal at −72 dBm andprovides a gain of 10 dBi to output the received signal at −62 dBm tothe interior circuit components.

Referring now to FIG. 56, there is illustrated the uplink signalstrengths when a power amplifier is located outside the window/wall5406. The transmitted signal has a strength of 18 dBm prior to reachingthe input of the horn antenna 5436. The antenna 5436 provides a gain of10 dBi to transmit the signal at 28 dBm. The window/wall 5406 causes anapproximately 40 dB total loss dropping the signal strength to −12 dB.The horn antenna 5438 provides a 10 dBi gain to the signal and outputsthe signal at −2 dBm. The power amplifier 5416 provides a 26 dB gain tooutput the signal at 24 dBm to the port 1 input of the circulator 5412.The power circulator 5412 provides a further 2 dB loss to output thesignal to the antenna 5410 at 22 dBm. The signal is transmitted from theantenna 5410 having a gain of 45 dB and a loss of 2 dB to provide atransmitted signal strength of 65 dBm.

Referring now to FIG. 57, there is illustrated the uplink signalstrengths when the power amplifier 5702 is located inside of thebuilding. The internal power amplifier 5702 is used when one needs morepower to be transmitted from the inside terminal. Prior to input to thepower amplifier 5702 the signal has a strength of 18 dBm within thebuilding. The power amplifier 5902 provides a 26 dB gain to transmit thesignal at 44 dBm to the input of the horn antenna 5436. The horn antenna5436 provides a 10 dBi gain and the transmitted RF signal is at 54 dBm.The transmitted signal experiences an approximately 40 dB loss throughthe window/wall 5404 that drops the signal strength to 14 dBm on theoutside portion of the window/wall 5404. The receiving horn antenna 5438provides a gain of 10 dBi to increase the signal strength to 24 dBm atthe output of the horn antenna 5438 that is provided to port 1 of thecirculator 5412. The circulator 5412 causes a 2 dB loss to drop thesignal strength to 22 dBm. The transmitting antenna 5410 provides afurther gain of 45 dB and loss off 2 dB to provide a transmitted outputsignal strength of 65 dBm.

Referring now to FIG. 58, there is illustrated the gains and losses onthe downlink when no power amplifier is included. A signal having a −103dBm strength is received by the antenna 5410. The antenna 5410 providesa gain of 45 dB and a loss of 2 dB. This provides a 60 DBM signal at theoutput of the antenna 5410 that is input to port 2 of the circulator5412. The circulator 5412 provides a further 2 dB loss to the signalproviding a −62 dBm signal from port 3 that is provided to the input ofthe horn antenna 5418 that provides a gain of the 20 dBi. A signalhaving a value of −42 dBm is transmitted from the horn antenna 5418through the window/wall 5406. The window/wall 5406 provides anapproximately 40 dB loss to the transmitted signal providing a −82 dBmsignal at the receiving horn antenna 5420. The horn antenna 5420provides a further 20 dBi gain to the signal that is output at −62 dBmto the remaining circuitry of the inside portion 5408 of the device.

Referring now to FIG. 59, there is illustrated the signal strengths atvarious points of an uplink when no power amplifier is provided. Thetransmitted signals are provided at a strength of 18 dBm to the input ofthe horn antenna 5432. The horn antenna 5432 provides a gain of 20 dBito output a signal at 38 dBm through the window/wall 5406. Thewindow/wall 5406 causes an approximately 40 dB loss to the signal suchthat the receiving horn antenna 5438 receives a signal at −2 dB. Thereceiving horn antenna 5438 boosts the signal to 18 dBm with a gain of20 dBi. The 18 dBm signal is input to port 1 of the circulator 5412. Thecirculator 5412 causes a 2 dB loss to the signal which is output throughport 2 at 60 dBm. The transmitting antenna has a gain of 45 dB and aloss of two dB to cause a transmitted signal from the antenna at 59 dBm.

The above described dB losses with respect to the window/wall andvarious system components are all approximate values. System includingother dB loss values and gain may also be used with respect to theembodiments described herein. It would be known to one skilled in theart of the manner for determining the dB losses that would be associatedwith a particular wall or window and the associated system components.One example of the manner in which the dB values may be determined isillustrated in U.S. Provisional Application No. 62/369,393, filed Aug.1, 2016, entitled REGENERATION, RETRANSMISSION OF MILLIMETER WAVES FORINDOOR PENETRATION, and U.S. Provisional Application No. 62/425,432,filed Nov. 22, 2016, entitled REGENERATION, RETRANSMISSION OF MILLIMETERWAVES FOR BUILDING PENETRATION USING HORN ANTENNAS, each of which isincorporated herein by reference.

Horn Antennas

Referring now to FIG. 60, there is illustrated a further alternativeembodiment using a horn antenna is used for the transmission of signalsthrough the window or wall. As before, the millimeter wave transmissionsystem 5402 includes an exterior portion 5404 located on an exterior ofa window or wall 5406 and interior portion 5408 located on the interiorof the wall or window. The exterior portion 5404 includes an antenna5410 for transmitting and receiving signals to an exterior source.

The transmitted and received signals are processed at a 28 GHzcirculator 5412. The port 3 signals are provided to the input of a poweramplifier 5414. Similarly, the output of a power amplifier 5416 isconnected to input port 1 such that signals to be transmitted areprovided to port 2 of the circulator 5412 for transmission by antenna5410. The signals output from the power amplifier 5414 are provided to a28 GHz horn antenna 5418. The horn antenna 5418 transmitted to the RFsignals provided from the power amplifier 5414 through the window orwall 5406 to a receiving horn antenna 5420. The receive signals areoutput from the horn antenna 5420 to a modulator circuit 5422 fordemodulation. The demodulator 5422 in addition to receiving the receivesignal from for an antenna 5420 receives a signal output from a phaselocked loop/local oscillator 5424. The phase locked loop/localoscillator 5424 is controlled responsive to a clock generation circuit5426. The demodulated signal is provided from the demodulator 5422 toanalog-to-digital converter 5428. The digital signal is routed via arouter 5432 the appropriate receiving party.

Signals to be transmitted are received from inside the building at therouter 5430. In a one embodiment this will comprise a Wi-Fi router. Therouter 5430 provides digital signals to a digital to analog converter5432 converts the signals into an analogue format. The analog signalsare then modulated by a modulator 5434. The modulator 5434 modulates thesignals responsive to input from the phase locked loop/local oscillator5424 under control of the clock generation circuit 5426. The modulatedsignals from modulator 5434 are output through the window/wall 5406through a horn antenna 5436. The signals transmitted by horn antenna5436 or received by a receiving horn antenna 5438 located on theoutside. The output of the horn antenna 5438 is provided to the inputpower amplifier 5416 that amplifies the signal for transmission from theantenna 5410 after passing through circulator 5412.

The horn antennas 5418, 5420, 5436 and 5438 can have high gains of up to20 dB. The antenna patterns of these antennas will have side lobes andfront lobes. The front lobes are projected toward a receiving antenna.In order to shield the surrounding environment from emissions from theside lobes of the horn antennas 5418, 5420, 5436 and 5438, shielding6202 may be added over the horn antennas to provide adequate protectionto the environment in the vicinity of the device. The shielding 6002 actas absorbers to block the signals from the surrounding environment andmay comprise any material required to contain and absorb the emissionsof the horn antennas to a localized area contained within the shieldingenclosure 6002.

Patch Antennas

Referring now to FIG. 61, there is illustrated an alternative embodimentusing patch antennas 6102 for transmissions of signals through a windowor wall 6104. The signals transmitted via the patch antennas 6102 areprocessed in one of the manners described herein above for transmittingsignals through a window or wall 6104. The patch antennas 6102 generatedirectional radio wave beams to tunnel through low-e glass or walls. Thedevice provides an optical or RF tunnel through the window or wall 6104without requiring the drilling of any holes or the creation of some typeof signal permeable portal. The millimeter wave transmission system 6100includes an exterior portion 6106 located on an exterior of a window orwall 6104 and interior portion 6108 located on the interior of thewindow or wall 6104. The exterior portion 6106 includes an antenna 6110for transmitting and receiving signals to an exterior source, such as abase station. Link budgets between the base station and the antenna mustbe satisfied. In a preferred embodiment, the antenna comprises a 28 Hzantenna. However, it will be realized by one skilled in the art thatother antenna operating bandwidths such as 24 GHz, 39 GHz, 60 GHz andother bandwidths may be utilized.

The transmitted and received signals that are received at the antenna6110 are provided from the interior 6108 are processed by transceiverprocessing circuitry 6112. The transceiver processing circuitry 6112 maycomprise any of the circuitries described herein above for placing thereceived signals at the antenna 6110 or signals received from the inside6108 of the building in order to enable their transmission through awindow or wall 6104 or convert from the format capable of passingthrough the window or wall 6104 for external transmission via antenna6110. The transceiver processing circuitry 6112 can down convertedfrequencies to lower frequency EM waves that can penetrate through glassand walls and also be amplified using antenna array. Components withinthe transceiver processing circuitry 6112 may comprise things such as,but are not limited to, an RF circulator, power amplifiers, up downconverters, RF transmission circuitries, optical transmissioncircuitries, etc.

The transceiver processing circuitry 6112 places the signals in a formatfor transmission through the window or wall 6104. The signals outputfrom the transceiver processing circuitry 6112 on line 6114 are providedto patch antenna 6102 a. The patch antenna 6102 a transmits the RF oroptical signals provided from the transceiver processing circuitry 6112through the window or wall 6104 to a receiving patch antenna 6102 b. Thepatch antennas 6102 may transmit/receive over a wide frequency band from24 GHz up to e-band. Within this range a particular band of operationsfor the patch antennas 6102 is utilized. These bands include, but arenot limited to, 24 GHz band; 28 GHz A1 band; 28 GHz B-1, A3 and B2bands; 31 GHz band; 39 GHz band; and 60 GHz band. The patch antennas6102 may be of different configurations to provide varying levels ofgain therefrom. In one embodiment the antennas may be configured toprovide 18 dB of gain.

The received signals are output from the patch antenna 6102 b on line6116 to transceiver processing circuitry 6118 for demodulation andfurther processing. The transceiver processing circuitry 6118 mayinclude any of the various configurations described herein above withrespect to the interior transceiver circuit. The demodulated andprocessed signal is provided from the transceiver processing circuitry6118 to a Wi-Fi router 6120 to be transmitted to receiving deviceswithin the structure.

Signals to be transmitted to an exterior receiver are received frominside the building at the Wi-Fi router 6120. The Wi-Fi router 6120provides signals to transceiver processing circuitry 6118 that convertsthe Wi-Fi data signals into an RF format that will transmit across thewall or window 6104 as discussed above. The RF signals are output fromthe transceiver processing circuitry on line 6120 to patch antenna 6102c. The modulated signals from patch antenna 6102 c are transmittedthrough the window/wall 6104. The signals transmitted by patch antenna6102 c are received by a receiving patch antenna 6102 d located on theexterior of the building. The output of the patch antenna 6102 d isprovided on line 6124 to the transceiver processing circuitry 6112. Thesignals are converted into the format that is needed to enabletransmissions of the signals from the antenna 6110. This format maycomprise 24 GHz, 28 GHz, 39 GHz, 60 GHz; current cellular LTEfrequencies; 3.5 GHz CBRS; 5 GHz; 24, 28, 39, 60, 70, 80 GHz mm-bands orany other frequency band suffering from signal loss issues whentransmitted through a window or wall.

Referring now to FIG. 62, there is provided an illustration of the patchantenna array 6202. The patch antenna array 6202 comprises a first layer6202 and a second layer 6204 located above the first layer 6202. Thefirst layer 6202 is connected directly to the window or wall 6104. Eachlayer 6202/6204 contains multiple patch antennas 6206. Each of the firstand second layers 6202/6204 transmit signals across the window or wall6104. The patch antenna array 6202 may transmit on all millimeter wavebands such as 24 GHz, 28 GHz, 39 GHz, 60 GHz, etc. The multiple patchantennas 6206 can be configured in rectangular, circular or ellipticalconfigurations to generate a directional beam for carrying a trafficpayload.

Referring now to FIG. 63, there is illustrated one of the patch antennas6206 of the patch antenna array 6202 of FIG. 62. The patch antenna 6206has an overall width along a first edge 6302 of 1.23 mm, and a length of1.56 mm on a second edge 6304. The patch antenna 6206 defines a slot6306 to which a transmission line 6308 connects to the patch antenna6206. The slot 6306 has a length of 0.36 mm along a first edge 6310 anda width of 0.1 mm on each side 6312 of the transmission line 6308. Thepatch antenna 6206 is generated on a substrate 6314 made of FR408. Thepatch antenna 6206 has a relative permittivity of 3.75, a loss tangentof 0.018 and a thickness of 0.127 mm.

Referring now to FIG. 64, there is illustrated a transmission beamsimulation for the antenna of FIG. 63 using HFSS (high-frequencystructure simulator). The single patch antenna generates a transmissionbeam 6402 as illustrated in FIG. 64 that has a peak gain of 3.8 dB and a3 dB beam width of 80°. Design and simulations of the patch antenna areperformed using ANSYS HFSS with microstrip feed structure to prepare formanufacturing. The sidelobe radiation can be absorbed with absorbingmaterial and the main lobe is directed towards the receiving unit.

By using a patch antenna array as illustrated in FIG. 62, a highlydirectional, high gain beam may be generated as generally illustrated inFIG. 65. A plurality of patch antennas 6502 within a patch antenna array6504 may each transmit an individual beam 6506. Each of the individualbeams 6506 have an associated directionality and gain. The output of thepatch antenna array 6504 will combine each of the individual patchantenna beams 6506 to create a combined array transmission beam 6508.The combined transmission beam 6508 will have a better directionalityand larger gain than each of the individual beam 6506 generated by theindividual patch antennas 6502. Thus, by generating the transmissionbeam using the patch antenna array 6504, a beam 6508 having sufficientgain and directionality to pass through a window or wall to a receiverand overcome associated signal losses will be possible.

Referring now to FIG. 66, there is illustrated a further embodiment of amicrostrip patch antenna array 6602 utilizing a microstrip antenna arrayfor 60 GHz bandwidth applications. The microstrip patch antenna array6602 comprises a 2×8 microstrip patch antenna array 6604 using aconductor-backed coplanar waveguide (CB-CPW) loop feed 6605. The patchantenna array 6604 consist of an upper substrate layer 6604 and a lowersubstrate layer 6606. The conductor backed coplanar waveguide 6605 islocated on the lower substrate layer 6606 comprising a 32 mm×28 mm planemade of quartz having a dielectric constant of 3.9, a loss tangent of0.0002 and a thickness of 0.525 mm. The plane 6606 defines an input 6610connecting to transmission lines that provides inputs to patch antennas6612 of a 2×8 patch antenna array defined on the plane 6606 that definethe CPW-fed loop. The upper substrate layer 6604 defines multiple patchantennas 6614 thereon on a Rogers RO3003 substrate having a thickness ofapproximately 0.254 mm, a dielectric constant of 3.00 and the losstangent of 0.001. This type of antenna provides a gain of 18 dB boardside at 61 GHz and has a bandwidth of approximately 57 GHz to 64 GHz.

Referring now more particularly to FIG. 67, there is illustrated a patchantenna element 6700. Multiple ones of these patch antenna elements 6700are located upon the multilayer patch antenna array as discussedhereinabove. The antenna element 6700 includes a patch 6702 having alength L and a width W. The patch 6702 is fed from an input transmissionline 6704 that is connected with a feed network and is resting upon asubstrate 6706 having a height h. The microstrip patch antenna includesa first radiating slot 6708 along a first edge of the patch 6702 and asecond radiating slot 6710 along a second edge of the patch 6702. Theelectronic field at the aperture of each slot can be decomposed into Xand Y components as illustrated in FIG. 68. The Y components are out ofphase and cancel out because of the half wavelength transmission line6704. The radiating fields can be determined by treating the antenna asan aperture 6800 as shown in FIG. 68 having a width W 6802 and a heighth 6804.

The transmission line model can be further analyzed in the followingmanner. G_(r) is the slot conductance and B_(r) is the slot susceptance.They may be determined according to the equations:

$G_{r} = \left\{ {{\begin{matrix}\frac{W^{2}}{90\;\lambda_{0}^{2}} & {{{for}\mspace{14mu} W} < \lambda_{0}} \\\frac{W}{120\lambda_{0}} & {{{for}\mspace{14mu} W} > \lambda_{0}}\end{matrix}B_{r}} = \frac{2\pi\;\Delta\;\ell\sqrt{ɛ_{eff}}}{\lambda_{0}Z_{0}}} \right.$

The input admittance of the patch antenna 6700 can be approximated as:

$Y_{in} = {Y_{slot} + {Y_{0}\frac{Y_{slot} + {j\; Y_{0}\;{\tan\left( {\beta\left( {L + {2\;\Delta\;\ell}} \right)} \right)}}}{Y_{0} + {j\; Y_{slot}{\tan\left( {\beta\left( {L + {2\Delta\;\ell}} \right)} \right)}}}}}$where A1 is the end effect of the microstrip.The rectangular patch antenna 6700 will resonate when the imaginary partof the input admittance goes to zero.

The end effect may be calculated according to the equation:

${\Delta\ell} = {0.412{h\left( \frac{ɛ_{eff} + {0.3}}{ɛ_{eff} - 0.258} \right)}\frac{\left( {W/h} \right) + 0.264}{\left( {W/h} \right) + 0.8}}$${L + {2{\Delta\ell}}} = {\frac{\lambda_{g}}{2} = \frac{\lambda_{0}}{2\sqrt{ɛ_{eff}}}}$$ɛ_{eff} = {\frac{ɛ_{r} + 1}{2} + {\frac{ɛ_{r} - 1}{2}\left( {1 + \frac{10h}{W}} \right)^{- 0.5}}}$

The resonant frequency of the patch antenna 6700 is given by:

$f_{r} = \frac{C}{2\sqrt{ɛ_{eff}}\left( {L + {2\Delta\;\ell}} \right)}$Typically the width W of the aperture is given by:

$W = {\frac{c}{2f_{r}}\left( \frac{ɛ_{r} + 1}{2} \right)^{{- 1}/2}}$

In addition to using patch antennas for generating highly directionaland highly gained beams for the transmission of signals through a windowor wall, patch antennas may be utilized using the application of orbitalangular momentum (OAM) to signals transmitted therethrough in order toincreased bandwidth on communication links between patch antennasthrough a window or wall. This is more fully illustrated in thefollowing descriptions beginning with FIG. 69.

FIG. 69-76 illustrates a multilayer patch antenna array 6902 that may beutilized for transmitting Laguerre-Gaussian (LG), Hermite-Gaussian (HG),Ince-Gaussian (IG), or orbital angular momentum (OAM) signals such asthose signals described in U.S. patent application Ser. No. 15/398,5611,filed on Jan. 4, 2017, entitled MODULATION AND MULTIPLE ACCESS TECHNIQUEUSING ORBITAL ANGULAR MOMENTUM, which is incorporated herein byreference in its entirety. The multilayer patch antenna array 6902includes a first antenna layer 6904 for transmitting a first orderedbeam, a second antenna layer 6906 for transmitting a second ordered beamand a third layer 6908 for transmitting a third ordered beam. Each ofthe layers 6904, 6906 and 6908 are stacked on a same center. While thepresent embodiment is illustrated with respect to a multilayer patchantenna array 6902 including only three layers, it should be realizedthat either more or less layers may be implemented in a similar fashionas described herein. On the surface of each of the layers 6904, 6906 and6908 are placed patch antennas 6910. Each of the patch antennas areplaced such that they are not obscured by the above layer. The layers6904, 6906 and 6908 are separated from each other by layer separatormembers 6912 that provide spacing between each of the layers 6904, 6906and 6908. The configuration of the layers of the patch antenna may be inrectangular, circular or elliptical configurations to generateHermite-Gaussian, Laguerre-Gaussian or Ince-Gaussian beams.

The patch antennas 6910 used within the multilayer patch antenna array6902 are made from FR408 (flame retardant 408) laminate that ismanufactured by Isola Global, of Chandler Ariz. and has a relativepermittivity of approximately 3.75. The antenna has an overall height of125 μm. The metal of the antenna is copper having a thickness ofapproximately 12 μm. The patch antenna is designed to have an operatingfrequency of 73 GHz and a free space wavelength of 4.1 mm. Thedimensions of the input 50 Ohm line of the antenna is 280 μm while theinput dimensions of the 100 Ohm line are 66 μm.

Each of the patch antennas 6910 are configured to transmit signals at apredetermined phase that is different from the phase of each of theother patch antenna 6910 on a same layer. Thus, as further illustratedin FIG. 71, there are four patch antenna elements 6910 included on alayer 6904. Each of the antenna elements 7504 have a separate phaseassociated there with as indicated in FIG. 71. These phases include π/2,2(π/2), 3(π/2) and 4(π/2). Similarly, as illustrated in FIG. 72 layer6906 includes eight different patch antenna elements 6910 including thephases π/2, 2(π/2), 3(π/2), 4(π/2), 5(π/2), 6(π/2), 7(π/2) and 8(π/2) asindicated. Finally, referring back to FIG. 69, there are included 12patch antenna elements 6910 on layer 6908. Each of these patch antennaelements 6910 have a phase assigned thereto in the manner indicated inFIG. 69. These phases include π/2, 2(π/2), 3(π/2), 4(π/2), 5(π/2),6(π/2), 7(π/2), 8(π/2), 9(π/2), 10(π/2), 11(π/2) and 12(π/2).

Each of the antenna layers 6904, 6906 and 6908 are connected to acoaxial end-launch connector 6916 to feed each layer of the multilayerpatch antenna array 6902. Each of the connectors 6916 are connected toreceive a separate signal that allows the transmission of a separateordered antenna beam in a manner similar to that illustrated in FIG. 70.The emitted beams are multiplexed together by the multilayered patchantenna array 6902. The orthogonal wavefronts transmitted from eachlayer of the multilayered patch antenna array 6902 in a spatial mannerto increase capacity as each wavefront will act as an independent Eigenchannel. The signals are multiplexed onto a single frequency andpropagate without interference or crosstalk between the multiplexedsignals. While the illustration with respect to FIG. 70 illustrates thetransmission of OAM beams at OAM 1, OAM 2 and OAM 3 ordered levels.

It should be understood that other types of Hermite Gaussian andLaguerre Gaussian beams can be transmitted using the multilayer patchantenna array 6902 illustrated. Hermite-Gaussian polynomials andLaguerre-Gaussian polynomials are examples of classical orthogonalpolynomial sequences, which are the Eigenstates of a quantum harmonicoscillator. However, it should be understood that other signals may alsobe used, for example orthogonal polynomials or functions such as Jacobipolynomials, Gegenbauer polynomials, Legendre polynomials and Chebyshevpolynomials. Legendre functions, Bessel functions, prolate spheroidalfunctions and Ince-Gaussian functions may also be used. Q-functions areanother class of functions that can be employed as a basis fororthogonal functions.

The feeding network 6918 illustrated on each of the layers 6904, 6906,6908 uses delay lines of differing lengths in order to establish thephase of each patch antenna element 6910. By configuring the phases asillustrated in FIGS. 69-72 the OAM beams of different orders aregenerated and multiplexed together.

Referring now to FIG. 73, there is illustrated a transmitter 7302 forgenerating a multiplexed beam for transmission. As discussed previously,the multilayered patch antenna array 6902 includes a connector 6916associated with each layer 6904, 6906, 6908 of the multilayer patchantenna array 6902. Each of these connectors 6916 are connected withsignal generation circuitry 7304. The signal generation circuitry 7304includes, in one embodiment, a 60 GHz local oscillator 7306 forgenerating a 60 GHz carrier signal. The signal generation circuit 7304may also work with other frequencies, such as 70/80 GHz. The 60 GHzsignal is output from the local oscillator 7306 to a power dividercircuit 7308 which separates the 60 GHz signal into three separatetransmission signals. Each of these separated transmission signals areprovided to an IQ mixer 7310 that are each connected to one of the layerinput connectors 6916. The IQ mixer circuits 7310 are connected to anassociated additive white gaussian noise circuit 7312 for inserting anoise element into the generated transmission signal. The AWG circuit7312 may also generate SuperQAM signals for insertion into thetransmission signals. The IQ mixer 7310 generates HG, LG, IG, OAMsignals in a manner such as that described in U.S. patent applicationSer. No. 14/323,082, filed on Jul. 3, 2014, now U.S. Pat. No. 9,331,875,issued on May 3, 2016, entitled SYSTEM AND METHOD FOR COMMUNICATIONUSING ORBITAL ANGULAR MOMENTUM WITH MULTIPLE LAYER OVERLAY MODULATION,which is incorporated herein by reference in its entirety.

Using the transmitter 7302 illustrated in FIG. 73. A multiplexed beam(Hermite Gaussian, Laguerre Gaussian, Ince Gaussian, etc.) can begenerated as illustrated in FIG. 74 at a specific frequency for highspeed tunneling. This type of mode division multiplexing (MDM) achieveshigher throughput with one frequency and multiple LG, HG or IG beams. Asillustrated, the multilayered patch antenna array 6902 will generate amultiplexed beam 7402 for transmission. In the present example, there isillustrated a multiplex OAM beam that has twists for various order OAMsignals in a manner similar to that disclosed in U.S. patent applicationSer. No. 14/323,082. An associated receiver detector would detect thevarious OAM rings 7404 as illustrated each of the rings associated witha separate OAM processed signal.

When signals are transmitted in free space (vacuum), the signals aretransmitted as plane waves. They may be represented as described hereinbelow. Free space comprises a nonconducting medium (σ=0) and thusJ=σE=0.

From experimental results Ampere's law and Faraday's law are representedas:

$\overset{\rightarrow}{B} = {{\mu\overset{\rightarrow}{H}\mspace{14mu}{\nabla x}\; H} = {{\frac{\partial D}{\partial t} + {J\mspace{14mu}{Ampere}^{\prime}s\overset{\rightarrow}{D}}} = {{\epsilon\;\overset{\rightarrow}{E}\overset{\rightarrow}{J}} = {{\sigma\overset{\rightarrow}{E}\mspace{14mu}{\nabla{xE}}} = {\frac{- {\partial B}}{\partial t}\mspace{14mu}{Faraday}^{\prime}s}}}}}$If there is propagation in the z direction and therefore E and H are inthe xy plane.

Without the loss of any generality E may be oriented in the x-directionand H may be oriented in the y-direction thus providing propogation inthe z-direction. From Ampere's-Maxwell equation, the following equationsare provided:

${\nabla{xH}} = {{\frac{\partial D}{\partial t}\mspace{14mu}{\nabla{xH}}} = {\begin{matrix}\overset{\hat{}}{x} & \overset{\hat{}}{y} & \overset{\hat{}}{z} \\\frac{\partial}{\partial x} & \frac{\partial}{\partial y} & \frac{\partial}{\partial z} \\H_{x} & H_{y} & H_{z}\end{matrix}}}$${\left( {\frac{{\partial H}z}{\partial y} - \frac{{\partial H}y}{\partial z}} \right)^{\overset{\hat{}}{x}} + \left( {\frac{{\partial H}z}{\partial z} - \frac{{\partial H}z}{\partial x}} \right)^{\overset{\hat{}}{y}} + \left( {\frac{{\partial H}y}{\partial x} - \frac{{\partial H}x}{\partial y}} \right)^{\overset{\hat{}}{z}}} = {\frac{\partial}{\partial t}\epsilon\; E}$

Next, the vectorial wave equations may be represented as:

$\mspace{20mu}{{\nabla{xH}} = {{\frac{\partial D}{\partial t} + {J\mspace{14mu}{\nabla{xH}}}} = {\epsilon\frac{\partial E}{\partial t}}}}$$\mspace{20mu}{{\nabla{xE}} = {{\frac{- {\partial B}}{\partial t}\mspace{14mu}{\nabla{xE}}} = {{- \mu}\frac{\partial H}{\partial t}}}}$  ∇xB = 0  ∇xE = S   ∇x∇xH = ∇(∇H) − ∇²H = −∇²H  ∇x∇xE = ∇(∇E) − ∇²E = −∇²E$\mspace{20mu}{{\nabla{x\left( {{\nabla x}H} \right)}} = {{\nabla{x\left( {\in \frac{\partial E}{\partial t}} \right)}} = {{\in {\frac{\partial}{\partial t}\left( {{\nabla x}E} \right)}} = {- {\in {\mu\frac{\partial}{\partial t}\left( {\frac{\partial}{\partial t}H} \right)}}}}}}$$\mspace{20mu}{{\nabla^{2}H} = {{+ \epsilon}\;\mu\frac{\partial^{2}}{\partial t^{2}}H}}$$\mspace{20mu}{{{\nabla^{2}H} - {{\epsilon\mu}\frac{\partial^{2}}{\partial t^{2}}H}} = {{0{\nabla{x\left( {{\nabla x}E} \right)}}} = {{\nabla{x\left( {{- \mu}\frac{\partial}{\partial t}H} \right)}} = {{{- \mu}\frac{\partial}{\partial t}\left( {{\nabla x}H} \right)} = {{{{- \mu}\frac{\partial}{\partial t}\left( {\epsilon\frac{\partial E}{\partial t}} \right)} + {\nabla^{2}E}} = {{+ {\mu\epsilon}}\frac{\partial^{2}}{\partial t^{2}}E}}}}}}$$\mspace{20mu}{{{\nabla^{2}E} - {{\mu\epsilon}\frac{\partial^{2}}{\partial t^{2}}E}} = 0}$

Therefore in general:{right arrow over (V)} ² {right arrow over (E)}+{right arrow over (K)} ²{right arrow over (E)}=0 E({right arrow over (r)},t){right arrow over (E)}(r,t)={right arrow over (E)}({right arrow over(r)})e ^(−jwt) e ^(jkz) Propagating in z-directionTherefore:

${{\left( {\frac{\partial^{2}}{\partial x^{2}} + {\frac{\partial^{2}}{\partial y^{2}}{+ \frac{\partial^{2}}{\partial z^{2}}}}} \right){\overset{\rightarrow}{E}\left( \overset{\rightarrow}{r} \right)}e^{- {jwt}}e^{jkz}} + {\frac{W^{2}}{y^{2}}{\overset{\rightarrow}{E}\left( \overset{\rightarrow}{r} \right)}e^{- {jwt}}e^{jkz}}} = 0$In free space

$w = {\frac{1}{\sqrt{\mu\epsilon}} = {\left. \rightarrow c \right. = {{\frac{1}{\sqrt{\mu\epsilon 0}}\mspace{14mu} k^{2}} = \frac{w^{2}}{c^{2}}}}}$Now:

${{\frac{\partial}{\partial z}{\overset{\rightarrow}{E}\left( \overset{\rightarrow}{r} \right)}}e^{jkz}} = {e^{jkz}\left\lbrack {\frac{\partial{\overset{\rightarrow}{E}\left( \overset{\rightarrow}{r} \right)}}{\partial z} + {jk{\overset{\rightarrow}{E}\left( \overset{\rightarrow}{r} \right)}}} \right\rbrack}$$\begin{matrix}{{\frac{\partial}{\partial z^{2}}{\overset{\rightarrow}{E}\left( \overset{\rightarrow}{r} \right)}e^{jkz}} = {{e^{jkz}\left\lbrack {\frac{\partial{\overset{\rightarrow}{E}\left( \overset{\rightarrow}{r} \right)}}{\partial z} + {jk{\overset{\rightarrow}{E}\left( \overset{\rightarrow}{r} \right)}}} \right\rbrack} + {e^{jkz}\left\lbrack {\frac{\partial^{2}{\overset{\rightarrow}{E}\left( \overset{\rightarrow}{r} \right)}}{\partial z^{2}} + {jk\frac{\partial{\overset{\rightarrow}{E}\left( \overset{\rightarrow}{r} \right)}}{\partial z}}} \right\rbrack}}} \\{= {{e^{jkz}\left\lbrack {{{jk}\frac{\partial\overset{\rightarrow}{E}}{\partial z}} - {k^{2}{\overset{\rightarrow}{E}\left( \overset{\rightarrow}{r} \right)}}} \right\rbrack} + {e^{jkz}\left\lbrack {\frac{\partial^{2}\overset{\rightarrow}{E}}{\partial z^{2}} + {jk\frac{\partial\overset{\rightarrow}{E}}{\partial z}}} \right\rbrack}}}\end{matrix}$Because

${{2k\frac{\partial E}{\partial z}}} ⪢ {\frac{\partial^{2}{E(r)}}{\partial z^{2}}}$Paraxial assumption

$\frac{{\partial^{2}{\overset{\rightarrow}{E}\left( \overset{\rightarrow}{r} \right)}}e^{jkz}}{\partial z^{2}} = {e^{jkz}\left\lbrack {{2{jk}\frac{\partial^{2}{\overset{\rightarrow}{E}\left( \overset{\rightarrow}{r} \right)}}{\partial z}} - {k^{2}{\overset{\rightarrow}{E}\left( \overset{\rightarrow}{r} \right)}}} \right\rbrack}$Then:

${\left( {\frac{\partial^{2}}{\partial x^{2}} + {{\frac{\partial^{2}}{\partial y^{2}}{+ 2}}jk\frac{\partial^{2}}{\partial z}}} \right){E\left( {x,y,z} \right)}} = 0$Which may be represented in cylindrical coordinates as:

${\frac{\partial^{2}}{\partial x^{2}}{+ \frac{\partial^{2}}{\partial y^{2}}}} = {{\frac{1}{q}\frac{\partial}{\partial q}\left( {q\frac{\partial}{\partial q}} \right)} + {\frac{1}{q^{2}}\frac{\partial^{2}}{\partial\Phi^{2}}}}$

This provides a paraxial wave equation in cylindrical coordinates:

${{\frac{1}{q}{\frac{\partial}{\partial q}\left( {q\frac{\partial}{\partial q}} \right)}{E\left( {q,\Phi,z} \right)}} + {\frac{1}{q^{2}}\frac{\partial^{2}}{\partial\Phi^{2}}{E\left( {q,\Phi,z} \right)}} + {2jk\frac{\partial E}{\partial z}\left( {q,\Phi,z} \right)}} = o$P(z), q(z) Then:$E_{0} \sim e^{- {j{\lbrack{p + {\frac{k}{2q}{({x^{2} + y^{2}})}}}\rbrack}}}$

In general E_(o) can rotate on the xy-plane and the wave stillpropagates in the z-direction.

${\frac{\partial q}{\partial z} = 1}{\frac{\partial P}{\partial z} = {- \frac{j}{q}}}$

-   -   q ˜Curvature of the phase front near the optical axis.        q ₂ =q ₁ +z        where q₂ is the output plane and q₁ is the input plane.∞∞

$\frac{1}{q} = {\frac{1}{R} - {j\frac{\lambda}{\pi\; W^{2}}}}$where

$\frac{1}{R}$is the curvature of the wavefront intersecting the z-axis.

Thus for a complete plane wave R=∞, the equation becomes:

${\frac{1}{q} = {\frac{1}{\left. R\rightarrow\infty \right.} - {j\frac{\lambda}{\pi\; W^{2}}}}}{q_{0} = {\frac{\pi\; W^{2}}{{- j}\lambda} = \frac{j\;\pi\; W_{0}^{2}}{\lambda}}}$where W_(o) is the beam waist.

${{q = {{q_{0} + z} = {\frac{j\;\pi\; W_{0}^{2}}{\lambda} + z}}}{w(z)}} = {w_{0}\sqrt{1 + \left( \frac{z}{z_{r}} \right)^{2}}}$${W^{2}(z)} = {W_{0}^{2}\left\lbrack {1 + \left( \frac{\lambda z}{\pi\; W_{0}^{2}} \right)^{2}} \right\rbrack}$${R(z)} = {z\left\lbrack {1 + \left( \frac{\pi\; W_{0}^{2}}{\lambda z} \right)^{2}} \right\rbrack}$${R(z)} = {z\left\lbrack {1 + \left( \frac{z_{R}}{z} \right)^{2}} \right\rbrack}$${\Phi(z)} = {\tan^{- 1}\left( \frac{z}{z_{R}} \right)}$$\theta = \frac{\lambda}{\pi\; w_{0}}$ z = z_(R)${w(z)} = {\sqrt{2}w_{0}}$

The Rayleigh length is:

$z_{R} = \frac{\pi\; n}{\lambda_{0}}$where n is the index of refraction.

${w_{0}^{2} = \frac{w^{2}}{1 + \left( \frac{\pi\; w^{2}}{\lambda R} \right)^{2}}}{z = \frac{R}{1 + \left( \frac{\lambda R}{\pi\; w^{2}} \right)^{2}}}$

The complex phase shift is represented by:

${{jP}(z)} = {{{Ln}\left\lbrack {1 - {j\left( \frac{\lambda z}{\pi\; w_{0}^{2}} \right)}} \right\rbrack} = {{{Ln}\;\sqrt{1 + \left( \frac{\lambda z}{\pi\; w_{0}^{2}} \right)^{2}}} - {j\;\tan^{- 1}\frac{\lambda z}{\pi\; w_{0}^{2}}}}}$

The real part of P(z) represents a phase shift difference between theGaussian beam and an ideal plane wave. Thus the fundamental mode isprovided:

${E_{0}\left( {x,y,z} \right)} = {{E_{0}\left( {r,z} \right)}\frac{w_{0}}{w}e^{- {j{({{jz} - \Phi})}}}e^{- {r^{2}{({\frac{1}{w^{2}} + \frac{jk}{2R}})}}}}$where: $\Phi = {\tan^{- 1}\frac{\lambda z}{\pi\; w_{0}^{2}}}$

Higher order modes may also provide other solutions. The solution ofrectangular equation:

${\left( {{\frac{\partial^{2}}{\partial x^{2}}{+ {\frac{\partial^{2}}{\partial y^{2}}{+ 2}}}}jk\frac{\partial}{\partial z}} \right){E\left( {x,y,z} \right)}} = 0$Can be determined in rectangular coordinates to be:

${E\left( {x,y,z} \right)} = {\sum_{mn}{C_{nm}E_{0}\frac{w_{0}}{w(z)}{H_{m}\left\lbrack \frac{\sqrt{2}x}{w(z)} \right\rbrack}{H_{n}\left\lbrack \frac{\sqrt{2}y}{w(z)} \right\rbrack}e^{\frac{- {({x^{2} + y^{2}})}}{{w{(t)}}^{2}}}e^{{- {j{({m + m + 1})}}}t\;{an}^{- 1}\frac{z}{z_{0}}}e^{j\frac{k{({x^{2} + y^{2}})}}{2{R{(z)}}}}}}$$\mspace{20mu}{z_{0} = \frac{kw_{0}^{2}}{2}}$$\mspace{20mu}{{w(z)} = {w_{0}\sqrt{1 + \frac{z^{2}}{z_{0}^{2}}}}}$  C₆₀ ⇒ TEM_(OD)$\mspace{20mu}{{R(z)} = {{z + \frac{z_{0}^{2}}{z}} = {{\frac{z_{0}^{2}}{z}\left( {1 + \frac{z^{2}}{z_{0}^{2}}} \right)} = {{\frac{z_{0}^{2}}{zw_{0}^{2}}{w^{2}(z)}} = {\frac{kz_{0}}{2z}{w^{2}(z)}}}}}}$

The solution of cylindrical coordinates of equation:

${{\frac{1}{\rho}\frac{\partial}{\partial\rho}\left( {\rho\frac{\partial}{\partial\rho}} \right){E\left( {\rho,\varnothing,z} \right)}} + {\frac{1}{\rho^{2}}\frac{{\partial^{\bigwedge}2}{E\left( {\rho,\varnothing,z} \right)}}{\delta\varnothing^{2}}} + {2jk\frac{\partial{E\left( {\rho,\varnothing,z} \right)}}{\partial z}}} = 0$Can be determined in cylindrical coordinates to be:

${E\left( {\rho,\varnothing,z} \right)} = {\sum\limits_{\ell\rho}{C_{\ell\rho}E_{0}\frac{w_{0}}{w(z)}\left( \frac{\sqrt{2}\rho}{w(z)} \right)^{\ell}{L_{\ell}^{\rho}\left( \frac{\sqrt{2}\rho}{w(z)} \right)}e^{- \frac{\rho^{2}}{w^{2}{(t)}}}e^{{- {j{({{2\rho} + \ell + 1})}}}{ta}\; n^{- 1}\frac{z}{z_{0}}}e^{j\;\ell\;\varnothing}e^{j\frac{k\rho^{2}}{2{R{(z)}}}}}}$The equation

$L_{\ell}^{\rho}\left( \frac{\sqrt{2}\rho}{w(z)} \right)$may also be shown as

${L_{\ell}^{\rho}\left\lbrack \frac{2\rho^{2}}{w^{2}(t)} \right\rbrack}.$

The lowest mode is the most important mode and in fact this transversemode is identical for both rectangular and cylindrical coordinates.

${\varphi\left( {\ell,{P;z}} \right)} = {\left( {{2P} + \ell + 1} \right)\tan^{- 1}\frac{z}{z_{0}}}$TEM₀₀^(rect) = TEM₀₀^(Cyl)C₀₀ = 1 H₀ = 1 L₀⁰ = 1 then$\left. {TEM}_{00}\Rightarrow{{E\left( {\rho,z} \right)} \sim {E_{0}\frac{w_{0}}{w(z)}e^{- \frac{\rho^{2}}{w^{2}{(t)}}}e^{{- {jtan}^{- 1}}\frac{z}{z_{0}}}e^{{jk}\;\frac{\rho^{2}}{2{R{(z)}}}}}} \right.$

Referring now to FIG. 75, there is illustrated a receiver 7502 fordemultiplexing signals received from a multiplexed signal generatedusing the transmitter 7302 of FIG. 73. The receiver 7502 includes amultilayer patch antenna array 7502 such as that described herein above.The multilayer patch antenna array 7502 receives the incomingmultiplexed signal 7504 and each layer 7504, 7506, 7508 of the antennaarray 7502 will extract a particular order of the received multiplexedsignal from each of the connector outputs 7516 of a particular layer.The signals from each of the connectors 7516 are applied to a mixercircuit 7506 that demultiplexes the received signal in a manner similarto that discussed with respect to U.S. patent application Ser. No.14/323,082 using a 60 GHz local oscillator signal from oscillator 7508.The demultiplexed signal may then be read using, for example, areal-time oscilloscope 7510 or other signal reading device. Each of thethree transmitted signals is thus decoded at the receiver 7502 that weretransmitted in each of the ordered OAM signals received from thetransmitters 602. In a further embodiment, a demultiplexing approachusing SPP (spiral phase plate) may also be applied to detect OAMchannels.

The signals transmitted by the transmitter 7302 or the receiver 7502 maybe used for the transmission of information between two locations in avariety of matters. These include there use in both front haulcommunications and back haul communications within a telecommunicationsor data network.

The multilayered patch antenna array 7502 may transmit both HermiteGaussian beams using the processing discussed with respect to U.S.patent application Ser. No. 14/323,082 or Laguerre Gaussian beams. Whentransmitting Laguerre Gaussian beams information may be transmitted in anumber of fashions. A spiral phase plate and beam splitter approach maybe used, a dual OAM mode antenna approach may be used or the patchedantenna described herein may be utilized. These implementations would bebeneficial in both fronthaul and backhaul applications.

In order to transmit several OAM modes of order l and amplitude a_(l)^(OAM), the antenna elements must be fed by an input signal according tothe equation:

${a_{n}^{feed}\frac{1}{\sqrt{N}}{\sum\limits_{l = 0}^{N - 1}{a_{l}^{OAM}e^{{- j}2\pi\frac{ln}{N}}}}},{n \in \left\{ {0,\ldots\mspace{14mu},{N - 1}} \right\}},$

Note that the number of elements in the multilayer patch antenna array7502 limits the number of possible OAM modes due to sampling. Due toaliasing, modes of order greater than N/2 are actually modes of negativeorders.

${b_{l^{\prime}}^{OAM} = {{\frac{1}{\sqrt{N}}{\sum\limits_{p = 0}^{N - 1}{b_{p}^{feed}e^{j\; 2\pi\frac{{pl}^{\prime}}{N,}}p}}} \in \left\{ {0,\ldots\mspace{14mu},{N - 1}} \right\}}},{h_{pn} = {\beta e^{- {jkr}_{np}}\frac{\lambda}{4\pi\; r_{np}}}},{r_{pn} = \sqrt{D^{2} + R_{t}^{2} + R_{r}^{2} - {2R_{t}R_{r}{\cos\left( \theta_{np} \right)}}}},{\theta_{pn} = {2{\pi\left( \frac{n - P}{N} \right)}}},{\beta = \sqrt{g_{t}g_{r}}}$Single Mode Link Budget

H_(tot) = U^(H)HUb^(OAM) = H_(tot)a^(OAM)${\frac{P_{r}}{P_{t}}(l)} = {{\frac{b_{l}^{OAM}}{a_{l}^{OAM}}}^{2} = {{\sum\limits_{p = 0}^{N - 1}{\sum\limits_{n = 0}^{N - 1}{\frac{\beta}{N}e^{{- j}l\theta_{np}}e^{{- j}{kr}_{np}}\frac{\lambda}{4\pi r_{np}}}}}}^{2}}$Asymptotic Formulation

The object is to determine an asymptotic formulation of the Link budgetat large distances, i.e. when D→+(∞), we seek the leading term for eachvalue of l Link budget −l are the same.

The link budget is asymptotically given by:

${\frac{P_{r}}{P_{t}}\left( {l} \right)} = {{\frac{\lambda\beta}{4\pi{{l}!}}\left( \frac{kR_{t}R_{r}}{2} \right)^{|l|}\frac{1}{D^{|l|{+ 1}}}}}^{2}$

From the Fraunhofer distance 2 (2max(R_(t),R_(r)))²/λ=200λ, the linkbudget asymptotically tends to straight lines of slope −20 (|l|+1) dBper decade, which is consistent with an attenuation in 1/D^(2|l|+2).

Asymptotic Expressions with Gains and Free Space Losses

Gains and free space losses may be determined by.

${\frac{P_{r}}{P_{t}}\left( {l} \right)} = {\frac{Ng_{t}}{{l}!}\left( \frac{4{\pi\left( {\pi\; R_{t}^{2}} \right)}}{\lambda^{2}} \right)^{|l|}\frac{Ng_{r}}{{l}!}\left( \frac{4{\pi\left( {\pi\; R_{t}^{2}} \right)}}{\lambda^{2}} \right)^{|l|}\left( \frac{\lambda}{4\pi\; D} \right)^{2|l|{+ 2}}}$${L_{FS_{eq}}(l)} = \left( \frac{4\pi D}{\lambda} \right)^{2|l|{+ 2}}$${G_{eq}(l)} = {\frac{Ng}{{l}!}\left( \frac{4{\pi\left( {\pi R^{2}} \right)}}{\lambda^{2}} \right)^{|l|}}$

For a fixed value of |l|, each equivalent gain increases R^(2|l|) Sothat the link budget improves by a factor of R^(4|l|). On the contrary,for a fixed value of R, when |l| increases, the link budget decreasessince asymptotically the effect of D is greater than those of R_(t) andR_(r).

Referring now to FIG. 76, there is illustrated a 3-D model of a singlerectangular patch antenna designed for 2.42 GHz and only one linearpolarization. The radiation pattern for this antenna is illustrated inFIG. 77.

FIG. 78a illustrates the radiation patterns of the circular array for anOAM mode order l=0 due to the higher grating lobes. FIGS. 78b, 78c and78d illustrate the radiation patterns for the OAM mode orders in l=0(FIG. 78b ), l=1 (FIG. 78c ), and l=2 (FIG. 78d ) in the vicinity of thearray axis.

Asymptotic OAM path loss is illustrated by:

${\frac{P_{r}}{P_{t}}\left( {l} \right)} = {\frac{Ng_{t}}{{l}!}\left( \frac{4{\pi\left( {\pi R_{t}^{2}} \right)}}{\lambda^{2}} \right)^{|l|}\frac{Ng_{r}}{{l}!}\left( \frac{4{\pi\left( {\pi R_{t}^{2}} \right)}}{\lambda^{2}} \right)^{|l|}\left( \frac{\lambda}{4\pi D} \right)^{2|l|{+ 2}}}$When assuming e-band frequencies, a distance of 1000 m and a reasonablepatch antenna element gains, other parameters may be calculatedincluding the diameter for the transmitter and receiver array rings,number of antennas, etc.

The production of the patch antennas 7510 are carried out through adesign and layout process as generally illustrated in FIG. 79, a cleanroom and lithography procedure for production of the antenna asgenerally illustrated in FIG. 80 and a final testing process asillustrated in FIG. 81. Referring now to FIG. 79, the design and layoutprocess is more particularly described. Initially, the patch antenna isdesigned and simulated at step 7902 using ANSYS HFSS with a microstripfeed structure. ANSYS HFSS comprises a high-frequency structuralsimulator. The software within the device stimulates 3-D full waveelectromagnetic field. The ANSYS HFSS creates a GDSII file (graphicdatabase system file used to control integrated circuit photomaskplotting) from the HFSS simulation and exports the GDSII file to an AWR(Applied Wave Research Corporation) Microwave Office (MWO) layout atstep 7904. In order to measure the antenna with ground signal groundprobe feeding, a previously design conductor backed coplanar waveguideto microstrip transition design that has been fabricated using AgilentMomentum is also imported at step 7906 as a GDSII Agilent Momentum fileinto the AWR MWO Layout. The two designs are brought together at step7908 and a weight and etch compensation of 12 μm is added to the lateraldimensions to account for isotropic wet etch used in the fabricationprocess. The final GDSII file for the layout is exported at step 7910and provided to a clean room for fabrication at step 1912.

Referring now to FIG. 80, there is illustrated the clean room processfor patterning a copper layer on the FR408 laminate. Initially, thedouble-sided Cu FR408 laminate is cut using scissors at step 8002 to anappropriate size (typically 1.5″×1.5″). The FR408 laminate is cleaned byrinsing the laminate at step 8004 with acetone, isopropanol (IPA) andnitrogen (N₂) and dried in a solvent hood or using program 2 of a CPKSolvent Spinner with the appropriate chuck. The laminate is dehydratebaked at 130° C. for two minutes on a hot plate (for example, a ColeParmer digital hotplate) at step 8006. Next, at step 8008,hexamethyldisilizane (HMDS) is deposited on the laminate by a rainmethod using a Yield Engineering YES—310 vacuum hood oven. The laminatesamples are placed in the HMDS oven for 20 minutes to improve resistadhesion. Next, at step 8010, the mask is cleaned using program 2 of aCPK Solvent Spinner with the appropriate chuck. The mask is furthercleaned using an automated mask cleaner (Ultratech Mask Cleaner) usingprogram 0 DIW only at step 8012.

The lithography process is performed at steps 8014-8034. First, ShipleyS1813 photoresist is spun on to the backside of the laminate at step8014 to protect the ground layer using for example a Brewer Science CeeSpin Coater System. In one embodiment, the spin coater system willoperate at 3000 rpm with 3000 rpm/s for 60 seconds. The sample is softbaked at step 8016 at 115° C. for 90 seconds on a hot plate and hardbaked at step 8018 at 130° C. for 60 seconds on the hotplate. S1813resist is spun onto the top side pattern copper layer at step 8022. Inone embodiment, this is carried out at 3000 rpm with 3000 rpm/s for 60seconds. The sample is soft baked at 115° C. for 90 seconds on a hotplate at step 8022. The top side of the sample is exposed at step 8024with 110 mJ/cm2 using Karl Suss MA6 BA6 Contact Aligner/Printer. Next,the circuit is developed at step 8026 with Microposit MF-319 for 60seconds in a beaker. The sample is rinsed with deionized water (DIW) andN₂ in a base hood. A reactive ion etching process is performed at step8032 to remove excess photoresist using Techniques Series 85 RIE. Thisis achieved by applying 02 only at 180 mTorr with 50 W for 15 seconds.The sample is hard baked at step 8034 at 130° C. for 60 seconds on a hotplate. The lithography is checked at step 8036 under a Leica Inm Opticalmicroscope to make sure the lithography is correct and that the gaps aredefined and not overdeveloped.

The 12 μm copper layer is etched at steps 8038-8046. The copper isetched in one minute intervals at step 8038 by agitating the sample in aCu etchant. Inquiry step 8040 determines if the Cu etching process iscomplete, and if not, the sample is rotated at step 8042 by 900 andreturns to agitate the sample within the Cu etchant at step 8038. Wheninquiry step 8040 determines that the Cu etching process is completedcontrol passes to step 8044 wherein the sample is rinsed with DIW and N₂and dried within a base hood. The sample is checked at inquiry step 8046using a microscope to determine if the Cu has been completely removed.If not, control passes back to step 8038 for further agitation withinthe Cu etchant. If all of the Cu has been removed control passes to thestripping of the photoresist process.

The stripping of the photoresist occurs by first rinsing the sample withacetone, IPA, DIW and N₂ and drying within a solvent hood or usingprogram 2 in CPK Solvent Spinner with the appropriate chuck. The sampleis dehydrate baked at step 8050 at 130° C. for five minutes on a hotplate. The etched laminate samples are examined at step 8052 under amicroscope to make sure that gaps are etched with no over etching ofareas within the sample.

The created patch antenna may be tested as illustrated in FIG. 81 toconfirm operation of the antenna. Initially, at step 8102, a DC test isperformed upon the antenna to make sure that the G-S-G feed is notshorted. An RF test is performed at step 8104 to measure the S₁₁-ReturnLoss across the frequency bands using Agilent VNA on Cascade M150 probestation. The radiation pattern of the antenna may then be measured atstep 8106 at the appropriate frequencies using a NSI spherical nearfield scanner.

In a further configuration patch antennas can be used in conjunctionwith horn antennas to overcome the 40 dB losses occurring through awindow or wall. The above describe embodiments would also be configuredto meet FCC and OSHA requirements. In addition to the techniquesdescribed herein above, other near field techniques can be used fortransmitting the information through the window or wall.

Transceiver Chipsets

Referring now to FIG. 82A there is illustrated an embodiment fortransmitting RF signals through a window or wall 8202 using an RFtransceiver chipset that transmits a frequency that will receive signalsfrom a base station 8204 at a frequency that will not penetrate thewindow/wall 8202 without significant signal losses. The base station8204 transmits wireless signals 8206 to a building transmissionpenetration system 8230. The building transmission penetration system8230 includes a first transceiver 8232 implementing a transmissionchipset for receiving the signals 8206 from the base station 8204. Thefirst transceiver 8232 is linked with a second transceiver 8212implementing the transmission chipset over the bidirectionaltransmission links 3236 for signals being transmitted into a structureand a transmission link 8238 for signals being transmitted to theexterior of the structure to a base station 8204.

The second transceiver 8234 located on the interior of a structurecommunicates with a Wi-Fi router 8220 over transmission line 8222 andreception line 8224. The Wi-Fi router 8220 communicates with wirelessdevices located within the structure. Transmission lines 8222 and 8224allow bidirectional communications between the Wi-Fi router 8220 andsecond transceiver 8218 in a similar manner that lines 8214 and 8216allow bidirectional communications between second transceiver 8234. Thechipsets implementing in the first transceiver 8232 and the secondtransceiver 8234 may receive any number of frequencies including, butnot limited to, 3.5 GHz, 24 GHz, 28 GHz, 39 GHz, 60 GHz, 71 GHz, and 81GHz from the base station for conversion to a format that will penetratethe window/wall 8202 into the interior of the building and from theexterior of the building. The signals may use any protocol including,but not limited to, 2G, 3G, 4G-LTE, 5G, 5G NR (New Radio) and WiGi.

Referring now to FIG. 82B, there is illustrated a more particularembodiment of the system of FIG. 82A for a system 8200 for transmitting60 GHz or other bandwidth wireless signals through a window or wall8202. In this embodiment, a Peraso chipset is used for enablingtransmissions within the system 8200. A base station 8204 transmits 60GHz wireless signals 8206 to a millimeter wave system 8208. Themillimeter wave system 8208 includes a first 60 GHz transceiver 8210implementing the Peraso chipset for receiving the 60 GHz signals 8206from the base station 8204. The first Peraso transceiver 8210 is linkedwith a second 60 GHz transceiver 8212 implementing the Peraso chipsetover the transmission connection 8214 for signals being transmitted intoa structure and a transmission line 8216 for signals being transmittedto the exterior of the structure to a base station 8204.

The second Peraso transceiver 8212 is located on an outside of a windowor wall 8202 and transmits wireless signals to a third 60 GHztransceiver 8218 implementing the Peraso chipset on the interior of thewindow or wall 8202. The third Peraso transceiver 8218 located on theinterior of a structure communicates with a Wi-Fi router 8220 overtransmission line 8222 and reception line 8224. The Wi-Fi router 8220communicates with wireless devices located within the structure.Transmission lines 8222 and 8224 allow bidirectional communicationsbetween the Wi-Fi router 8220 and third Peraso transceiver 8218 in asimilar manner that lines 8214 and 8216 allow bidirectionalcommunications between second Peraso transceiver 8212 and first Perasotransceiver 8210. For TDD, typically 3 time slots would be assigned forTX and 1 time slot for RX, and therefore the slots would not collide asthey would be separated in time. Therefore, for two way communications,there are no issues in terms of interference at the same frequency. Forsituations where the same frequency and time are used, Full Duplexisolation using OAM twisted beams can be used where the TX is done with+1 helicity and RX is done with −1 helicity.

Referring now to FIG. 83, there is illustrated a further Peraso chipsetimplementation. FIG. 83 illustrates a repeater configuration wherein abase station 8302 communicates with a 60 GHz transceiver 8304implementing the Peraso chipset over a 60 GHz wireless communicationslink 8306 signals are bidirectionally transmitted between the Perasotransceiver 8304 to a second 60 GHz transceiver 8308 implementing thePeraso chipset. The second Peraso transceiver 2908 has a wireless 60 GHzcommunications link 8310 with a third 60 GHz transceiver 8312 that alsoimplements the Peraso chipset over a distance indicated generally by8314. The repeater 8316 consisting of Peraso transceiver 8304 and Perasotransceiver 8308 enable signals from the base station 8302 to be boostedand transmitted over larger distances to the Peraso transceiver 8312.The Peraso transceiver 8312 bidirectionally communicates with the router8318 over the communications lines 8322 and 8324. The repeaterconfiguration such as that described herein above can be used to extendthe range of transmission of the 60 GHz signal transmitted from the basestation 8302.

FIG. 84A illustrates a top level block diagram of a Peraso transceiver8460 that may be used for transmissions as described hereinabove. A pairof antennas 8462 are used for transmitting (8460B) and receiving (8462A)60 GHz signals. Received signals according to one of the embodimentsdescribed hereinabove are passed from antenna 8462A to a demodulator8464 wherein the signals received from the antenna 8462A are demodulatedresponsive to signals provided from the phase locked loop/localoscillator block 8466. The demodulated signals are passed to an analogto digital converter 8468 for the analog signals to be converted to adigital format responsive to the demodulated signal and a clock signalprovided by clock generator 8470. The digital signal is provided at anoutput 8472.

Signals to be transmitted are provided at input 8474 in a digital formatand converted from digital to analog format at the digital to analogueconverter 8476 responsive to a clock signal from clock generator 8470.The analog signal is modulated within modulator 8478 responsive to theanalog signal and control signals from the phase locked loop/localoscillator block 8466. The modulated signals are transmitted fromantenna 8462B in one of the configurations described hereinabove fromthe Peraso transceiver 8460. The Peraso chipset is more particularlydescribed in the Peraso W110 WiGig Chipset Product Brief dated Dec. 18,2015 which is incorporated herein by reference.

Referring now to FIG. 84B, there is illustrated a more detailedapplication diagram of the Peraso chipset. While the Peraso chipset inthe 60 GHz band has been described, it will be realized by one skilledin the art that the chipset may utilize any frequency where the repeaterenables extension of signal transmission capabilities. Examples include,but are not limited to, millimeter bands, 28 GHz band, 39 GHz band, 2.5GHz band, CBRS band (3.5 GHz) and Wi-Fi band (5 GHz). The Peraso chipsetcomprises the W110 chipset that is targeted for use with WiGigapplications. The Peraso chipset employs a PRS1125 integrated circuit8402 and PRS4001 integrated circuit 8404 to implement the IEEE 802.11adfunctionality. The Peraso chipset implements a complete superspeed USB3.0 to WiGig solution. The PRS4001 low power WiGig baseband integratedcircuit 8402 incorporates the analog front end 8406 including digital toanalogue converters 8408, analog-to-digital converters 8410 and a phaselocked loop 8412. The PRS 4001 circuit 8402 further includes thebaseband physical layer 8414, Mac layer 8416 and two RISC CPU cores. ThePRS4001 circuit 8402 is IEEE 802.11ad compliant. A USB 2.0 and 3.0interfaces 8424 enable USB communications. The PRS4001 circuit 8402supports seamless connection to all Peraso 60 GHz radios.

The PRS1125 integrated circuit 8404 is a single chip direct conversionRF transceiver providing 60 GHz single ended receiver and transmitinterfaces. The PRS1125 circuit 8404 provides a transmit output power ofup to 14 dBm, better than −21 dB transmit EVM (16-QAM), receiver noiseless than 5 dB and a receiver conversion gain of greater than 70 dB.Integrated single ended 60 GHz antenna interfaces include a transmitdata path 8418 and a received data path 8420. A phase locked loop 8422tunes to all channels of IEEE 802.11ad using an integrated controller.The Peraso chipset provides for wireless storage, wireless display andmulti-gigabyte mobile wireless applications. The antennas 8426 compriseNA graded patch antennas with 8.5 dBi gain across the entire 60 GHzband.

Communications between Peraso chipset transceivers may be carried out ina number of fashions in order to control throughput therebetween. Asillustrated in FIG. 85, communications between the first Perasotransceiver 8502 and a second Peraso transceiver 8504 may be carried outin series over a single communications channel 8506. In this case, thedata is transmitted serially one item after the other over the singlecommunications channel 8506. FIG. 86 illustrates a parallel transmissionconfiguration. In this configuration, transmissions between transceiver8502 and transceiver 8504 occur over multiple channels 8608 operating inparallel. In this configuration, different data streams may betransmitted at the same time over the parallel communication channels8508 in order to increase data throughput. In the parallelconfiguration, a data stream is petitioned in two multiple sub streamsand sent on the separate parallel channels 8508. The results may then becombined together at the receiver 8504.

FIG. 87 illustrates a functional block diagram of a Peraso transmitter8702 located on an exterior side of a window or wall 8704. Since thePeraso transceiver 8702 is located on the exterior of the window or wall8704 some manner for providing power to the Peraso transceiver 8702 isneeded. A power unit 8706 located on the external side of the wall 8704may power the Peraso transceiver 8702 in a number of fashions. In oneimplementation, the power unit may comprise solar cells and solargeneration circuits for generating power. In one embodiment, the maximumpower consumption for the Peraso transmitter located on the externalwall or window is 15 W. In order for the transceiver to provide atransmit power of 14 dBm or approximately 25 mW, 15 W of consumed poweris created. If the 15 W of consumed power is required for 20 hours aday, approximately 300 Whrs of energy are needed to support thetransceiver each day. A power unit having an efficiency of 1.25operating for 24 hours can provide approximately 375 Whrs of energy. The375 Whrs are divided by 3.5 (the approximate number of sun hours in thewinter) to provide a needed solar capacity for the transceiver of 100 W.

Another method for providing power to an exterior Peraso transmitter isillustrated in FIG. 88. A laser 8802 located on the interior is used totransmit energy within a laser beam 8804 to photodiodes 8806 located onthe external Peraso transmitter. The laser beam 8804 would betransmitted through a window since a wall would block the beam. Therequired power for the transmitted laser beam is defined by:

${{P_{Optic} = \frac{P_{Electric}}{{Eff}_{Optics} \times {{Eff}_{PVCells}(\eta)}}}Q{E\left( {Eff}_{{PV} - {Cell}} \right)}},{\eta = {{\frac{R_{\lambda}}{\lambda} \times \frac{hc}{e}} \approx {\frac{R_{\lambda}}{\lambda_{\mu m}} \times {1.2}4}}}$$\eta = {R\frac{{1.2}4}{\lambda_{\mu m}}}$

The efficiency of the optics Eff_(Optics) varies depending upon the typeof glass that is being transmitted through. Window glass may be of acommercial or residential nature. For residential window glasses such asClimaGuard 70/36, the optics efficiency is 0.68 at a transmissionwavelength of 445 nm. For commercial window glass such as SunGuard SN 68the optics efficiency is 0.64 at a transmission wavelength of 445 nm.

The efficiency of the silicon photodiodes Eff_(PV Cells) is defined by:

${\eta\left( {Eff}_{{PV} - {Cell}} \right)} = {{R\frac{1.24}{\lambda_{\mu m}}} = {0.69}}$

Thus, the optical power needed to be transmitted at 450 nm can bedetermined using the optics and photodiode efficiencies in the followingmanner:

$P_{Optic} = {\frac{P_{Electric}}{{Eff}_{Optics} \times {{Eff}_{PVCells}(\eta)}} = {\frac{15}{{0.6}4 \times {0.6}9} = {34\mspace{14mu} W}}}$Therefore, the number of laser diodes needed to provide 34 W of powerwith 450 nm, 4.5 W blue diodes would be

$\frac{45\mspace{14mu} W}{4.5\mspace{14mu} W}$or approximately 8 diodes.VCSEL Alignment and Power

Referring now to FIG. 89, there is illustrated a VCSEL 8902. Since oneVCSEL is located on the outside of a window and a second VCSEL islocated on the inside of the window, there must be some manner foraligning the optical transmission links that are provided from one VCSELto the other. One manner in which this alignment may be achieved is byhaving alignment holes 8904 located at multiple positions on the VCSEL8902. In the embodiment illustrated in FIG. 89, the alignment holes 8904are located at each corner of the VCSEL 8902. These alignment holes 8904are used in the manner illustrated in FIG. 90 to align a first VCSEL8902 a with a second VCSEL 8902 b. Thus, by visually aligning each ofthe alignment holes 8904 located at each corner of the VCSEL 8902 a andVCSEL 8902 b, the optical transmission circuitry within the VCSELs maybe aligned through the window 9002.

Rather than using the external power inputs, the VCSEL 8902 located on awindow may be powered using other methods as illustrated in FIG. 91,FIG. 91 illustrates a VCSEL 9102 on an interior of a window or wall 9002and a VCSEL 9106 located on an exterior of a window or wall. Power 9108is provided directly to the internal VCSEL 9102 via some type of inputconnection. A power coupling device 9110 within the internal VCSEL 9102couples with a similar power coupling device 9112 within the externalVCSEL 9106. If the VCSEL's 9102 and 9106 are located on a transparentwindow, a photo inductor or other type of optical power coupler may beutilized for power coupling devices 9110 and 9112. If the VCSEL's 9102and 9106 are located on opposite sides of an opaque wall, or windowblocking optical signals inductive coupling devices such as coil anddoctors may be used for power coupling devices 9110, 9112. In thismanner, the power coupling devices 9110 provides power to the powercoupling device 9112 to power the external VCSEL 9106.

System Power

Referring now to FIG. 92, there is illustrated the manner in which powermay be provided to the external system component 9202 located within theexternal portion of the system and the internal system components 9204located within the internal portion 5608. The internal system component9204 comprises the antennas, modulator, demodulator and other componentsdiscussed herein above for generating signals for transmission anddetermining signals that have been received. The external systemcomponents 9202 consist of the circulator, power amplifiers and hornantennas described above. The internal system component 9204 isconnected to an internal power system 9206 that may plug into theelectrical power system located within the building. Since the internalsystem component 9204 and external system component 9202 are separatedby a window/wall 9002, there must be some manner for transmitting orproviding power to the external system components. One manner for doingso involves the use of a power system 9208 that is powered by a numberof solar panels 9210 that are located on the exterior of the building towhich the external system component 9202 are connected.

The power required from the power system 9208 to the external systemcomponents 9202 is approximately 0.76 W. One manner for providing this0.76 W power is through the use of solar panels 9210. Solar panelsproviding 0.76 W or 1 W may be utilized for the solar panels 9210. Withrespect to a 0.76 W power provision system, 0.76 W for 24 hours wouldrequire 18.24 W hours of power. If 18.24 W hours are provided at anefficiency of 1.25%, this will require 22.8 W hours. If an efficiency of22.8 W hours is divided by 3.5 hours (#number of daylight hours inwinter), a total result of 6.52 W is provided. Similarly for a 1 Wsystem, 1 W provided for 1 day requires 24 W hours. 24 W hours at a1.25% efficiency requires 30 W hours. 30 W hours divided by 3.5 hours ofsun available in the winter provides 8.57 W hours. The solar panels 9210used for providing power may be similar to those solar panels used forcharging smart phones and tablets. These type of panels include both 7 Wcharging panels and 9 W charging panels that meet the 0.76 W and 1 Wenergy levels requirements.

7 W portable solar chargers having high efficiency solar charging panelsnormally have a weight of 0.8 pounds. These devices have generaldimensions of 12.8×7.5×1.4 inches (32.5×19×3.5 cm). Other 7 W amorphoussolar power battery charger panels have a size of 15.8×12.5 ×0.8 inches(40×31.75×2 cm) and a weight of 3 pounds. Alternative 9 W chargingpanels with monocrystalline cells have dimensions ranging from8.7×10×0.2 inches (22×25.5×0.5 cm) and flexible solar panels have a sizeof 12×40 inches (30.5×100 cm). Other 9 W high-efficiency solar panelshave sizes from 8.8×12.2×0.2 inches (22.35×31×0.5 cm).

Referring now to FIG. 93, rather than utilizing solar panels, theexternal system components 9202 may utilize transmitted laser power forpowering the external system components rather than utilizing a solarpowered system. The internal system components 9204 have a power system9302 that provides power for all components on the interior portion of awindow or wall 9304. The power system 9302 has an internal powerconnection 9306 to for example, a power outlet located within thebuilding. The power system 9302 provides system power to the internalsystem components 9204 in a known manner. Additionally, the power system9302 provides power to a laser transmitter 9308. The laser transmittermay in one embodiment comprise laser diodes. The laser transmitter 9308generates a laser beam 9910 that is transmitted through a window 9304 toa photovoltaic receiver (PV receiver) 9312 located on the outside of thewindow 9304. The laser transmitter 9308 includes a set of optics todefine the beam size that is to be transmitted to the PV receiver 9312.The generated laser power may be defined according to the followingequations:

$P_{Optic} = \frac{P_{Electric}}{{Eff}_{Optics} \times {{Eff}_{PVCells}(\eta)}}$${Q{E\left( {Eff}_{{PV} - {Cell}} \right)}},{\eta = {{\frac{R_{\lambda}}{\lambda} \times \frac{hc}{e}} \approx {\frac{R_{\lambda}}{\lambda_{\mu m}} \times {1.2}4}}}$$\eta = {R\frac{{1.2}4}{\lambda_{\mu m}}}$

The optical power needed by the PV receiver that detects energy at 445nm may be defined in the following manner:λ=445 nmThis is the wavelength of the receiver laser.

R = 0.25  (Hamamatsu  Si − photodiode)$\eta = {{R\frac{1{.24}}{\lambda_{\mu m}}} = {0.69}}$Eff_(Optics) = 0.64  (Efficiency  of  Optics)$P_{Optic} = {\frac{P_{Electric}}{{Eff}_{Optics} \times {{Eff}_{{PV}\;{Cells}}(\eta)}_{Optic}} = {\frac{{0.7}6}{{0.6}4 \times {0.6}9} = {1.72\mspace{14mu} W}}}$Thus, in order to provide power at 445 nm a 2 W laser diode is needed.The PV receiver 9312 converts received laser light energy back intoelectricity. Power generated by the PV receiver 9312 responsive to thereceived laser beam 9310 is provided to the power system 9314. The powersystem 9314 and provides power to the external system component 9202 toenable their operation.

Referring now to FIG. 94, there is illustrated a further manner forpowering exterior components from an interior power source usinginductive coupling rather than utilizing solar panels or a laser source,the external system components 9202 may utilize power provided bymagnetic inductive or magnetic resonance coupling to the internal powersource through the window/wall 9404 for powering the external systemcomponents. The internal system components 9204 have a power system 9402that provides power for all components on the interior portion of awindow or wall 9404. The power system 9402 has an internal powerconnection 9406 to for example, a power outlet located within thebuilding. The power system 9402 provides system power to the internalsystem components 9204 in a known manner. Additionally, the power system9402 provides power to an inductive coil or magnetic resonator 9408. Theinductive coil or magnetic resonator 9408 enables a magnetic connectionwith a second inductive coil or magnetic resonator 9412 located on theexterior of the window/wall 9404. The inductive coils or magneticresonators 9408 and 9412 enable the inductive or resonate coupling ofpower from the internal power system 9402 to the external power system9414. Power received at the inductive coil or magnetic resonator 9412responsive to the received electromagnetic energy 9410 is provided tothe power system 9414. The power system 9414 and provides power to theexternal system component 9202 to enable their operation to transmitsignals through the window/wall 9404.

Also, in addition to the actively powered devices illustrated in FIGS.92, 93 and 94, a passively powered device may be used that provides nopowering to the exterior components but provides a shorter distance orhigher power from the internal components within the building.

The inductive coils 9408/9412 provide for inductive coupling of powerbetween the internal and external circuitries while the magneticresonators 9408/9412 use magnetic resonance coupling to transfer thepower between the circuitries. With respect to the inductive coils, thecoupling coefficient between the coils can be calculated in thefollowing manner. Referring now to FIG. 95, the mutual inductancebetween two circular loops separated by a distance d 9502 each with aradii of a 9504 and b 9506 can be calculated using Neumann's equation:

$M = {\frac{\mu}{4\pi}{\int{\int{\frac{\cos\;\epsilon}{r}{dsds}^{\prime}}}}}$where ds and ds' the incremental sections of the circular filaments andr is the distance between the two sections, which are defined as:r=√{square root over (a ² +b ² +d ²−2a cos(ø−ø′))},∈=ø−ø′, ds=adø, ds=bdø′

The substitution of the above into Neumann's equation results in:

$M = {\frac{\mu}{4\pi}\underset{0}{\overset{2\;\pi}{\int\int}}\frac{{ab}\mspace{11mu}{\cos\left( {\phi - \phi^{\prime}} \right)}}{a^{2} + b^{2} + d^{2} - {2a\mspace{11mu}{\cos\left( {\phi - \phi^{\prime}} \right)}}}d\;\phi\; d\;\phi^{\prime}}$

The integral of the above equation can be rewritten using ellipticalintegrals, yielding:

${M(m)} = {\frac{2\mu\sqrt{ab}}{m}\left\lbrack {{\left( {1 - \frac{m^{2}}{2}} \right){K(m)}} - {E(m)}} \right\rbrack}$where K(m) and E(m) are the elliptical integrals of first and secondkind, respectively, and m is defined as:

$m = \sqrt{\frac{4ab}{\left( {a + b} \right)^{2} + d^{2}}}$assuming values between 0 and 1.

The solutions of the elliptical integrals of the first and second kindcan be approximated using the following equations:

${{K(m)} = {\frac{\pi}{2} + {\frac{\pi}{8}\frac{m^{2}}{1 - m^{2}}}}}{{E(m)} = {\frac{\pi}{2} - {\frac{\pi}{8}m^{2}}}}$For low values of m, the power series representation shows reasonableaccuracy. However, as m increases both ellipticals diverge from thenumeric integration values. For the lip to inner goal of the first kind,as in approaches the unity, the solution asymptotically tends toinfinity much faster than a solution calculated by numeric integration.

The substitution of equation of K(m) and E(m) into the equation for M(m)yields:

${M(m)} = {\frac{\mu\pi\sqrt{ab}}{8}\frac{m^{3}}{1 - m^{2}}}$

Next, substituting the equation for m into the expression above resultsin the expression for the mutual inductance as a function of distancebetween two circular coaxial loops:

$M = \frac{\mu\pi a^{2}b^{2}}{\sqrt{\left( {a + b} \right)^{2} + {d^{2}\left\lbrack {\left( {a - b} \right)^{2} + d^{2}} \right\rbrack}}}$

For two coils with n_(1,2) turns, the expression can be adjusted,yielding:

$M = \frac{\mu\pi n_{1}n_{2}a^{2}b^{2}}{\sqrt{\left( {a + b} \right)^{2} + {d^{2}\left\lbrack {\left( {a - b} \right)^{2} + d^{2}} \right\rbrack}}}$which expresses the mutual inductance of two coils with n_(1,2) as afunction of distanced, the magnetic permeability of the materialsurrounding the coils p and the inner radius of the two coils.

Figure-of-Merit (U) can be described in terms of the Q factor, whichdescribes the ratio between the energy stored by loops of an inductorand the power dissipated in a given cycle. The Figure-of-Merit isdependent upon different coil parameters such as wire radius Ra, loopradius a, permeability of free space μ₀ (since the core of the loop isair), conductivity of the core material and d distance between primaryand secondary loops according to the equation:

$u = \frac{\sqrt{2\mu_{0}\omega\sigma}\pi\;{na}^{3}R_{a}}{d^{2}\sqrt{{4a^{2}} + d^{2}}}$

In one embodiment, the transmission coil would have the characteristicsof a loop radius of 6.25 cm, a wire radius of 10.25×10⁻³, 4 coil turns,a distance between the primary and secondary loops of 46 mm and anoperating frequency of 6.78 MHz.

Referring now to FIG. 96, there is illustrated a table providinginformation relating to the efficiencies of a coil over various coilradius, various number of turns within the coil and differing coilheights.

Referring now to FIGS. 97 and 98, there are illustrated the manner inwhich coils 9702 inductively coupled with each other and in whichresonator circuits 9802 inductively resonate with each other. FIG. 97illustrates how alternating current is provided from an oscillator 9704responsive to an input voltage 9706 to the L1 coil 9702. The alternatingcurrent within the L1 coil 9702 generates an alternating magnetic field9708 which in turn induces an alternating current in the secondary coilL2. This causes a current to be provided to rectifier 9710 that isprovided to a load 9712. The magnetic field generated by the primarycoil 9702 radiates approximately equally in all directions. The fluxescreated by the magnetic field drop rapidly with distance in accordancewith the inverse square law. Therefore, the secondary coil L2 9702 mustbe placed as close as possible to the primary coil L1 9702 to interceptthe most magnetic flux.

Referring now to FIG. 98, in order to overcome the major drawback ofinductive wireless charging caused by the requirement to closely couplethe coils thus demanding precise alignment and close proximity betweenthe coils, magnetic resonance wireless charging may be utilized.Magnetic resonance may be used for charging any active component frominside a building to the outside of the building by using differentsized coils. The basic concept behind magnetic resonance power transferis to tunnel energy from one coil to another in a directed fashionacross the window or wall instead of spreading energy omni-directionallyfrom the main coil. The magnetic resonance wireless charging circuitreceives an input voltage Vs across inputs 9804 that applies the voltageVs to oscillator 9806. The output of the oscillator 9806 is passedthrough a drive coil 9808. The drive coil 9808 generates a current inthe primary resonator circuit 9802 that includes a coil 9810 having acapacitor 9812 connected across the coil 9810. The resonator circuit9802 a couples with resonator circuit 9802 b to provide magneticresonance wireless charging. Resonator circuit 9802 b includes a coil9814 having a capacitor 9816 connected across the coil. The resonatorcircuit 9802 b couples with a drive coil 9818 connected to rectifier9820 this is used for driving a load 9822. The basic concept behindmagnetic resonator power transfer is that the energy from resonatorcircuit 9802 a is tunneled to resonator circuit 9802 b instead ofspreading omnidirectional from the primary coil 9802 a.

In order to use inductive coupling and magnetic resonance coupling toprovide for wireless power transfer from an interior of a building to anexterior of a building through a window or wall using the abovedescribed millimeter wave transmission system, differing designconsiderations must be dealt with depending upon whether inductivecoupling or magnetic resonance coupling is utilized. In order to providewireless power transfer using inductive coupling, a high magneticcoupling is required necessitating the distance between the transmittingpower unit and the receiving power unit being very small. Standardsavailable for inductive coupling wireless power transfer include Qi andPMA. Using the standards between 5 W and 15 W may be transmitted oversmall distances of 5-10 mm.

Wireless power transfer using magnetic resonance coupling, also referredto as highly resonant wireless power transfer (HR-WPA), uses a looselycoupled magnetic resonance for power transfer. High-quality factormagnetic resonators enable efficient energy transfer at lower couplingrates enabling power transfer over greater distances between thetransmitting and receiving power units while providing more positionalfreedom. Existing standards include Rezence (WiTricity) and WiPower(Qualcom).

Referring now to FIG. 99, there is illustrated a functional blockdiagram of a magnetic resonance wireless power transfer system which maybe utilized to power the millimeter wave system of the presentdisclosure. An AC voltage signal is provided at an AC input 9902. The ACvoltage signal is applied to an AC/DC converter 9904 that converts thealternating current signal into a direct-current signal. Thedirect-current signal from the AC to DC converter 9904 is applied to aDC/RF amplifier 9906. The DC/RF amplifier 9906 is a high-efficiencyswitching amplifier that converts the DC voltage into an RF voltagewaveform used for driving the source resonator. The RF voltage waveformfrom the DC/RF amplifier 9906 is applied to add impedance matchingnetwork 9908. The impedance matching network 9908 provides impedancematching and improves system efficiency. The signal from the impedancematching network 9908 is provided to the transmission side sourceresonator 9910 which links the signal to the receiver side deviceresonator 9912. The source resonator 9910 and the device resonator 9912are highly quality factor resonators that enable efficient energytransfer at lower coupling rates (greater distance and/or positionalfreedom) between the transmission side and receiver side located onopposite sides of a window or wall. This energy coupling is referred toas highly resonant wireless power transfer (HR-WPT). The powertransferred to the device resonator 9912 goes to a second impedancematching network 9914 and to a RF/DC rectifier 9916. The rectifier 9916is used for loads 9918 requiring a DC voltage and converts the receivedAC power back into a DC signal.

The source resonator 9910 and device resonator 9912 have characteristicsthat can be described by two fundamental parameters, namely, a resonantfrequency, ω₀, and an intrinsic loss rate, Γ. The ratio of these twoparameters defines the quality factor (Q) of the resonator, (Q=ω₀/2Γ) ameasure of how well the resonator stores energy. A resonator energyoscillates at the resonant frequency between the inductor (energy storedin the magnetic field) and the capacitor (energy stored in the electricfield) and is dissipated in the resistor. The resonant frequency and thequality factor of the resonator are defined as:

$\omega_{0} = \frac{1}{\sqrt{LC}}$$Q = {\frac{\omega_{0}}{2\;\Gamma} = {{\sqrt{\frac{L}{C}}\frac{1}{R}} = \frac{\omega_{0}L}{R}}}$

The expression for Q shows that decreasing the loss in the circuit,i.e., reducing R, increases the quality factor of the system. High-Qelectromagnetic resonators are typically made from conductors andcomponents with low absorption and as a result have relatively narrowresonant frequency widths.

By locating the source resonator 9910 in close proximity to the deviceresonator 9912 coupling may be achieved between the devices enabling theresonators to exchange energy. A schematic representation of coupledresonators is illustrated generally in FIG. 100. The source voltage is asinusoidal voltage source 10002 with an amplitude V_(g) at frequency ωwith equivalent generator resistance R_(g) 10004. The source and deviceresonator coils are represented by the inductors L_(S) 10006 and L_(D)10008 which couple through the mutual inductance M where M=k√{squareroot over (L_(S)L_(D))}. Each coil has a capacitor to forma resonator(C_(S) 10010 and C_(D) 10012). The resistances R_(S) 10014 and R_(D)10016 are parasitic resistances that include both ohmic and radiativelosses of the coils 10006, 10008 and resonant capacitors 10010 and 10012of the respective resonators. The load is represented by R_(L) 10018.

Analysis of the circuit of FIG. 100 provides the power delivered to theload resistor 10018 divided by the maximum power available from thesource in both the source and device are resonant at ω according to theequation:

$\frac{P_{L}}{P_{g,{{ma}\; x}}} = \frac{{4 \cdot U^{2}}\frac{R_{g}}{R_{s}}\frac{R_{L}}{R_{d}}}{\left( {{\left( {1 + \frac{R_{g}}{R_{s}}} \right)\left( {1 + \frac{R_{L}}{R_{d}}} \right)} + U^{2}} \right)^{2}}$$U = {\frac{\omega\; M}{\sqrt{R_{s}R_{d}}} = \frac{k}{\sqrt{\Gamma_{s}\Gamma_{d}}}}$where U is the figure-of-merit for the system.

The generator resistances 10014, 10016 and load resistance 10018 arechosen to provide the best system performance (done by the impedancematching network) in accordance with:

$\frac{R_{g}}{R_{s}} = {\frac{R_{L}}{R_{d}} = \sqrt{1 + U^{2}}}$The efficiency of the power transmission as defined above is thenmaximized in accordance with:

$\eta_{opt} = \frac{U^{2}}{\left( {1 + \sqrt{1 + U^{2}}} \right)^{2}}$

The best possible efficiency of a wireless power transfer system dependson the system figure-of-merit, which can be written in terms of themagnetic coupling coefficient between the resonators, k, and theunloaded resonator quality factors, Q_(S) and Q_(D).

$U = {\frac{\omega\; M}{\sqrt{R_{s}R_{d}}} = {k\sqrt{Q_{s}Q_{d}}}}$

Magnetic coupling coefficient (k) is a function of the relative sizes ofthe resonators, the distance between the resonators and the relativeorientation of the resonators. The above equation illustrates that usinghigh quality factor resonators allows for efficient operation even atlower coupling rates. This eliminates the need for precise positioningbetween the source and device resonators and provides for a greaterdistance between coils and more positional freedom and freedom ofmovement. The elimination of the need for precise positioning allows fora consumer to install the internal and external transceivers located onthe interior and the exterior of the window or wall.

The Figure-of-Merit U depends on different coil parameters such as wireradius Ra, loop radius a, permeability of free space μ_(o), the distanced between the primary and secondary loop and conductivity of the corematerial. Figure-of-Merit U can be expressed in terms of the Q-factor,which describes the ratio between the energy stored by loops and thepower dissipated in a given cycle.

$U = {{kQ} = \frac{\omega_{0}M}{R}}$${U\left( {\omega_{0},a,d,R_{a},\sigma} \right)} = \frac{\omega_{0}{M\left( {a,d} \right)}}{R\left( {a,\omega_{0},R_{a},\sigma} \right)}$m = 4a²/(4a² + d²)${K(m)} = {\frac{\pi}{2} + {\frac{\pi}{8}\frac{m^{2}}{1 - m^{2}}}}$${E(m)} = {\frac{\pi}{2} - {\frac{\pi}{8}m^{2}}}$$R_{r\;{ad}} = \frac{{\mu_{0}\left( {\omega_{0}a} \right)}^{4}\pi}{12\left( c^{3} \right)}$$R_{rohm} = {\sqrt{\frac{\mu_{0}\omega_{0}}{2\;\sigma}}\frac{a}{R_{\alpha}}}$R = R_(T) = R_(R) = R_(r ad) + R_(ohm)where σ denotes the kind of committee of the material and c is the speedof light.

Referring now to FIG. 101, there is provided a circuit diagram of apower generator for converting 50 Hz grid AC to kHz such as the DC/RFamplifier 9906 of FIG. 99. This illustrates a potential power source forthe wireless energy transfer system that utilizes a rectifying andswitching network to convert the power grid AC to the frequency ofoperation of the energy transfer system. FIG. 101 illustrates a simpleexample of a power source including a rectifier 10102 including fourdiodes 10104 and a switching network 10104 including four power MOSFETtransistors 10108. A capacitor 10110 is connected between the rectifier10102 and the switching network 10106. The resistance of the powersource is in the range of 250 m to 400 m. An input to the power sourceis provided across the rectifier 10102 across terminals 10112. Theoutput vi 10114 from the switching network 10106 is approximately asquare voltage. The Fourier components of a normalized Square signalf(t) is:

${{f(t)} = {\frac{4}{\pi}{\sum\limits_{{n = 1},3,5,\ldots}^{\infty}{\frac{1}{n}{\sin\left( \frac{2\; n\;\pi\; t}{T} \right)}}}}},$

Referring now to FIG. 102, there is illustrated the manner in whichImpedance matching may be utilized to overcome losses between resonators10202 and 10204. A schematic representation of the resonators 10202 and10204 are provided in the manner described previously for transmissionthrough a window 10206. Two resistors R_(thin) 10208 are inserted acrossthe inductors L_(S) and L_(D) respectively to mimic and model the Eddycurrent losses due to the thin silver layer in Low-e class. With properimpedance matching through the resistors and/or matching control usingthe previously describe impedance matching network, coils and resistor,the losses may be overcome by modifying the turns in the coil, the areaof the coils, the permeability (material type) of the coil, as well asthe frequency of the resonant frequency.

FIGS. 103 and 104 illustrate a perspective view and a side view of theoutside transmission circuitry 10302 and inside transmission circuitry10304 using the Peraso chipset and inductive or resonant coupling fortransmitting power from the inside transmission circuitry to the outsidetransmission circuitry. Outside transmission circuitry 10302 consists ofthe antenna 10306 that receives millimeter wave transmissions from abase station or other external transmission source. In an alternativeembodiment, the antenna 10306 may also comprise a Peraso transceiver toenable direct reception of transmissions from another Perasotransceiver. A Peraso transceiver 10308 is used for transmitting signalsthrough a window or wall separating the outside transmission circuitry10302 from the inside transmission circuitry 10304. A coil 10310 is usedfor inductive power transmission or magnetic resonance powertransmission from the interior of the building in the manner describedhereinabove. A circuit board 10312 is used for interconnecting theelectronic components of the antenna 10306, Peraso transceiver 10308 andthe coil 10310.

The inside transmission circuitry 10304 includes a Peraso transceiver10312 for transmitting and receiving signals with the Peraso transceiver10308 in the outside transmission circuitry 10302. An interior coil10314 enables inductive or magnetic resonance power coupling with theoutside transmission circuitry 10302. Additionally, a circuit board10316 enables interconnection between the Peraso transceiver 10312, thecoil 10314 and any other interior electronic circuitries.

With respect to window glass through which signals or power must betransmitted, the relative permittivity, power transmission, phase andreflection may be calculated according to the Drude model as shown bythe following:

$\epsilon_{r} = {1 - \frac{\omega_{p}^{2}}{\omega\left( {\omega + {i\gamma}} \right)}}$

-   -   ω_(p): Bulk plasma frequency    -   γ: intraband damping term    -   For silver: ω_(p)=9.6 ev, γ=0.0228 ev    -   ∈_(r)=(n+ik)²    -   k∝amount of loss due to absorption    -   Absorbed power:

$1 - {\left( e^{k2\pi f\frac{x}{c}} \right)\hat{}2}$Phase:

$\frac{2\pi nx}{c}$The values of ε_(r), n, k, absorb power and absorption loss areillustrated in FIG. 105.

The reflection loss for one layer may be defined as −10 log (1−R²) whilethe reflection loss for two layers may be defined as −10 log(1−2R²−R⁴−2R³). The absorption loss is defined as −20 log e^(−αx).Values for these are more particularly illustrated in FIGS. 106 107.These values may be determined based upon the reflectivity R:

$R = {\frac{1 - \sqrt{\epsilon_{r}}}{1 + \sqrt{\epsilon_{r}}}}^{2}$

And the absorption coefficient:

$\alpha = {\frac{2\omega}{c}k}$Application with Residential IP Network System

Current broadband systems use wired broadband with fiber connections totransmit information from the network provider to consumers. Forexample, AT&T U-verse has fiber to the node and copper to premises or insome cases provide fiber all the way to the premises. Fiber to thepremises systems are expensive and require a great deal of time todeploy. Other solutions are DirecTV, DLS modem from Frontier, and acable box of Charter or Comcast. Another solution has been theimplementation of a wireless delivery of broadband. However, whendelivering broadband using wireless, high-frequency RF waves, issuesarise with respect to signals that cannot penetrate through window glassand walls of homes and buildings.

In traditional cable TV or satellite networks using broadcast RF videotechnology, all content constantly flows downstream to each customer,and the customer switches the content at the set-top box. The customercan select from among many choices provided by the cable or satelliteprovider, that are provided via the pipe flowing into the home/business.The broadcast network is only one way of transmitting data from theprovider to the consumer. Thus far, the approach has been to placeantennas on the roof to receive the signals from a hub and then drillingthrough different floors to enable the signals to penetrate into thebuilding. This approach of delivery from the roof of the building toindividual units within the building is very costly and time-consumingfor operators. Another approach is to direct beams from the hub toindividual units but this may cause the signal to hit a window or wallof the building. The losses are introduced by the wall or window whenthe radio beams try to penetrate into the building. These losses arehuge for millimeter wave radio signals and therefore methods forproviding broadband delivery utilizing the above described techniqueswould be greatly beneficial.

One manner for overcoming the above-noted issues with respect towireless broadband transmissions is illustrated in FIG. 108. Bycombining existing residential IP network systems 10802 with millimeterwave transmission systems 10804 an improved combined residential IPnetwork system 10806 may be provided. The millimeter wave transmissionsystem 10804 has the advantage of higher bit rates, more precise beamforming and steering and smaller footprint components. Residential ITPnetwork systems 10802 comprise combination services consisting ofInternet, TV, and VoIP phone services. These services can be ordered ina bundle or separately and not all combination of services may beavailable. The TV services are based on IPTV (Internet protocoltelevision) used to deliver TV services. A network systems 10802 alsoutilize IP technology (Internet protocol technology) such that TV,computer, home phone and wireless devices are integrated to worktogether using the Internet protocol. This provides many usefulfeatures, more control over devices in the manner of delivery of theservices. The use of IP technology also provides for morepersonalization such that services may be tailored toward the exactneeds of the consumer. An example of this type of service is AT&TU-verse, DirecTV, DSL modem from Frontier, cable box of Charter orComcast. The residential IP network system 10802 and video backbonedelivers high-quality video, advanced functionality and otherapplications. The residential IP network system 10802 is a two-way IPnetwork provided to the customer's home via fiber to the premises orfiber to the node technology.

The millimeter wave system 10804 enables the transmission of signalsthrough a window or wall as was more fully described hereinabove. Bycombining the millimeter wave system 10804 with the residential IPnetwork system 10802, wireless broadband transmissions may be providedfrom a network provider to user devices located on the interior of thebuilding without losses occurring by transmissions of the signalsthrough a window or wall degrading system performance. Within thecombined residential IP system 10806, content will remain in the networkand only be provided to the customer when requested. Within the combinedresidential IP system 10806, the IP network is two-way. Switched videodelivery is not limited by the size of the “pipe” into thehome/business. The network allows for delivery of more content andfunctionality. The network creates the potential to provide customersmore choices, including niche programming of interest to diverseaudiences and more high definition (HD) programming.

Compared to “traditional” cable or satellite TV, a combined system 10806providing IPTV is a different, improved configuration enabling moreflexibility and creativity within the network. A combined system 10806using IPTV enables two-way interactivity versus a traditional one-waycable or satellite broadcast network. The two-way residential IP networkenables viewers to have more options to interact, personalize andcontrol their viewing experience. IP technology also allows for moreflexibility within a home network. With the combined system residentialIP network, all system receivers with any home or business are connectedon the same high-speed network. This enables one to connect gamingconsoles, laptops and other devices to the premises residential IPnetwork.

Watching IPTV on a combined system 10806 is different than streamingvideos over the public Internet. With IPTV, program is carried over anetwork providers residential IP network which allows the networkprovider to control video quality and the reliability of the service.Best effort Internet video can be subjected to delays due to a lowerbandwidth, high-traffic or poor connection quality. Since IPTV enablesTVs to communicate with other services, integrated high-speedInternet-based content and features may be brought to the TV screen. Forexample, online photos uploaded to the public or personal clouds can beseen directly from the TV.

Referring now to FIG. 109, there is more particularly illustrated afunctional block diagram of the residential IP combined system 10806 ofFIG. 108. Network content 10902 is provided from a service provider to amillimeter wave transmission system 10904. The network content 10902 maycomprise video, audio, Internet web pages or any other network basedmaterial. The millimeter wave system 10904 may operate at a number ofwavelengths in accordance with the systems described hereinabove withrespect to the transmission of signals from an exterior of the buildingto an interior of a building and from an interior of the building to anexterior of the building. The millimeter wave system 10904 would includeall of the various systems for transmitting bidirectionally between theinside and outside of the building described hereinabove. The millimeterwave system 10904 transmits broadband data to a residential IP system10906 located on the interior of a building. The millimeter wave system10904 can be on both sides of the glass or wall, allowing tunneling ofradio waves either via optics or RF. The millimeter wave system 10904 isconnected to a residential gateway 10906 directly via electronicintegration at the window unit. In an alternative embodiment, themillimeter wave system 10904 is wirelessly connected to a residentialgateway 10906 either on licensed band or unlicensed Wifi with beamforming. The residential IP system 1106 provides the broadband contentto a number of home devices 10908 located on the interior of thebuilding via wireline connections 10910 and wireless connections 10912.

FIG. 110 illustrates a functional block diagram of a residential IPnetwork system 11002. An input 11004 from a millimeter wave transmissionsystem that enables the transmission of millimeter waves from anexterior transmission unit to the interior of the structure providesbroadband signals to a residential IP network gateway 11006. Theresidential IP network gateway 11006 determines where the signal comingfrom the input 11004 needs to be routed and provides the output on oneof a plurality of possible outputs to the appropriate destination IPaddress associated with the device requesting the broadband information.The output lines may comprise a coaxial cable 11008, an ethernet cable11010 or existing phone line 11012. The coaxial cables 11008 may provideinputs to a set top box 11014 that then provides an output to a livingroom TV 11016 through for example an HDMI connection 11018. A firstethernet connection 11010 may connect to a set top box/DVR 11020. Afurther ethernet connection 11022 provides data to a second television11024. Ethernet connections 11010 may also provide data to a PC 11026 ora network drive 11028. The existing phone line connection 11012 would beprovided to a phone outlet 11030 for connection of a telephone. Finally,a Wi-Fi antenna 11032 provides the ability for the residential IPnetwork Gateway 110062 to provide a Wi-Fi network connection within astructure. The Wi-Fi network connection enables devices such as a PC11034, laptop 11036, iPad 11038 or iPhone 11040 to wirelessly connect tothe residential IP network Gateway 11006 to receive broadband data.

FIG. 111 illustrates the manner in which a millimeter wave system may beutilized to transmit information to a residential IP network system. Anaccess unit 11102 located on the outside of a structure wirelesslytransmits broadband data to CPE (customer premises equipment) units11104 located within one or more structures associated with theresidential IP network system. The access unit 11102 may receive thebroadband data for transmission to the CPE units 11104 via wirelesstransmissions or a hardwired connection. The wireless access providedbetween the access unit 11102 and the CPE units 11104 may be provided inany of a number of frequency bands including, but not limited tomillimeter bands 24 GHz, 28 GHz, 39 GHz, 60 GHz as well as 2.5 GHz, theCBRS band 3.5 GHz, Wi-Fi bands at 2.4 and 5 GHz. The signals aretransmitted from outside the structure to inside the structure using anyof the above described transmission techniques for transmitting signalsthrough a wall or window. Within the structure the CPE unit 11104 usesWi-Fi or other unlicensed bands within the premises to transmit signalsto Internet of things (IOT) devices 11106, PCs 11108, IP TVs 11110,closed circuit televisions 11112, IP telephones 11114 and Wi-Fiextenders 11116. These are only some examples of IP-based devices andany type of Wi-Fi connectable device may be utilized within thestructure for communications with the CPE 11104. The manner in whichbroadband data may be transmitted from the exterior of the structure tothe interior of the structure may be configured utilizing the abovedescribed millimeter wave transmission systems in a number of fashions.

FIG. 112 illustrates a first embodiment wherein the access unit 11102wirelessly transmits the broadband data to a millimeter wave systemtransceiver 11202 located on an external side of a window or wall 11204.The system is consumer installed with the repeater (transceiver 11202)outside of the building and a transceiver 11206 on the inside of thebuilding. This configuration uses millimeter wave transmitters on bothsides of the glass or wall enabling tunneling of radio waves usingeither optical signals or RF signals. The broadband signals areconnected directly to the CPE device 11104 via electronic integration atan integrated window unit to provide access to the residential IPnetwork 11208. The wireless transmissions to the millimeter wavetransceiver 11202 may be within any frequency band including, but notlimited to, millimeter wave bands such as to 24 GHz, 28 GHz, 39 GHz, 60GHz and 2.5 GHz; CBRS bands such as 3.5 GHz; and Wi-Fi bands such as 2.4and 5 GHz. Millimeter wave transceiver 11202 transmits the signalsthrough the window or wall 11204 to a second millimeter wave transceiver11206 located on the interior of the structure. The composition of themillimeter wave transceivers 11202 and 11206 may be any of thosediscussed herein above with respect to systems for transmitting signalsthrough a window or wall 11204. The interior millimeter wave transceiver11206 outputs received data to (or receives data from) a customerpremises equipment 11104 associated with the residential network IP11208. The millimeter wave transceiver 11206 and CPE 11104 may compriseintegrated equipment within a same box or device for receiving thesignals from the millimeter wave transceiver 11202 located on theexternal of the structure and providing the data to the residential IPnetwork 11208 and associated devices.

FIG. 113 illustrates an alternative embodiment wherein the access unit11102 wirelessly transmits the broadband data signals to the externalmillimeter wave transceiver 11202 as described previously with respectto FIG. 112. In this embodiment, millimeter wave transceivers areprovided on sides of the window or wall 11204 enabling tunneling ofradio waves using either optical signals or RF signals. The signalstransmitted through the window or wall 11204 are then wirelesslyconnected to the CPE 11104 using either an unlicensed band or unlicensedWi-Fi with beamforming. The external millimeter wave transceiver 11202transmits the data through the window or wall 11204 as described hereinto an internal millimeter wave transceiver 11206. The internalmillimeter wave transceiver 11206 incorporates a beamforming device orWi-Fi router that allows for transmission of the received signals usingbeam forming license or Wi-Fi to an integrated millimeter wavetransceiver 12002 and CPE 11104. The CPE 11104 transmits the data to theresidential IP network 11208 and associated devices.

Referring now to FIG. 114 there is illustrated a further embodiment of asystem for transmitting the broadband signals to a residential IPnetwork 11208 wherein the access unit 11102 wirelessly transmits thesignals to a millimeter wave transceiver 11402 located on an externalside of a window 11404 of a building or structure. A millimeter wavetransceiver 11402 is located on the outside of a window glass and useshigh power phased array and beamforming circuitry 11403 to enabletunneling of radio waves to wirelessly connect to the CPE 11408 locateda distance from the window 11404 using either a licensed band orunlicensed Wi-Fi. The millimeter wave transceiver 11402 includes ahigh-powered phased array 11403 providing beamforming or Wi-Fi routercapabilities for transmitting signals wirelessly through the window11404 to a millimeter wave transceiver 11406 located on an interior ofthe structure but placed at a location that is not directly on theopposite side of the window 11404. The millimeter wave transceiver 11406is integrated with the CPE 11408 that transmits the broadband data tothe residential IP network 11208 and associated devices.

The described system provides an optical or RF tunnel that allowssignals to be transmitted from outside a building to devices within thebuilding. Once the broadband access is delivered into the premises(residential or commercial), other unlicensed bands can be used insidethe premises. The optical or RF tunnel can also be used to allow signalsfrom the Internet of Things devices located within the building to gofrom inside to outside. In addition to the techniques described hereinabove, other near field techniques can be used for transmitting theinformation through the window or wall.

Millimeter Wave with Optical Networks

One of the challenges faced in the next generation broadband access atgigabyte rates is the need for running fiber to a home or business. Withfixed millimeter wave 5G wireless access technology existing opticalnetwork units (ONU) which are passive optical network (PON) end pointsmay be used for the aggregation of self-installed fixed wireless accesspoints. FIG. 115 illustrates a combination of a millimeter wave system11502 with optical data transfer systems 11504 such as GPON/NGPON2/vOLTHA. This combined system enables the control of bandwidthallocation from OLT to millimeter wave RUs as will be more fullydescribed herein below. Each of these optical data transfer systems11504 provide a manner for transmitting data from a central networklocation to the millimeter wave system 115022 that enables transmissionof the data in an RF format over the last drop (last 100 m) to a userpremises such as a home or business. The millimeter wave system 11502may use millimeter wave beamforming and beam steering technologies toensure quality of service to home applications in response todynamically changing network conditions. The millimeter wave system11502 provides connection to a residential gateway 11506 (such as the IPnetwork Gateway described hereinabove) to provide services to a userlocated within a home or business. The millimeter wave system 11502greatly increases the number of enterprise and residential buildingswhere 5G millimeter waves can be used to deliver Gigabit Ethernet. Thus,the millimeter wave system 11502 uses TDMA and SDN-controlled beamsteering for wireless last drop access to the millimeter wave system ata structure that transmits to a residential gateway 11506.

Since the residential gateway 11506 does not have the ability todynamically trigger or adjust dataflow operations between the millimeterwave system 11502 and the optical data transfer system 11504 based onnetwork conditions, the hybrid ONU and millimeter wave remote units asdescribed hereinbelow may implement innovative SDN enabled beam steeringmechanisms to achieve high quality experience with dynamic networkslicing mechanisms and optimized OLT-ONU signaling frameworks. Thus, asmore particularly illustrated in FIG. 116, the optical network dataflow11602 within the GPON/NG PON2/vOLTHA network 11504 and the dataflow11604 of the RF network 11502 may be balanced by the control system11606 such that the RF network 11502 provides sufficient resources tosupport the required optical network data flow 11602, and the opticalnetwork 11504 provides sufficient resources to support the RF networkdataflow 11604. The configuration involves configuring network deviceswithin the optical network 11504 and RF network 11502. Thus, the opticalnetwork 11504 and RF network 11502 are comprised of components which maybe remotely reconfigured by a central controller in order to enablebalancing of loads being transmitted by the networks. If sufficientresources are not present on either of the networks, networkconfiguration 11608 may be altered such that the network dataflowbetween the optical network 11504 and the RF network 11502 are balancedto prevent any bottleneck at the interface between the two networks. Thenetwork reconfiguration may utilize the network configuration controltechniques described in U.S. patent application Ser. No. 15/664,764,entitled ULTRA-BROADBAND VIRTUALIZED TELECOM AND INTERNET, filed on Jul.31, 2017, which is incorporated herein by reference.

The optical data transfer systems 11504 (FIG. 15) include GPON, NG PON2,vOLTHA or similar types of systems. Referring now to FIG. 117, withinGPON (gigabyte passive optical network) there are two main active typesof transmission equipment that are used, OLT (Optical Line Terminal)11702 and ONU (Optical Network Unit) 11704 or ONT(Optical NetworkTerminal) 11706. The Optical Line Terminal 11702 is in a central office11708 and controls the information going both directions across theOptical Distribution Network. The OLT 11702 contains CSM (Control andSwitch Module), ELM (EPON Link Module, PON card), and redundancyprotection. The Optical Network Unit 11704 converts the optical signalstransmitted through fiber 11710 from the OLT 11702 to electrical signalsthat are sent to individual subscribers at the customer premises 11712.The ONU 11704 can also send data coming from the subscriber to the OLT11702. The Optical Network Terminal 11706 is essentially the same as theONU 11704. ONT 11706 is an ITU-T (ITU Telecommunication StandardizationSector) term and ONU 11704 is an IEEE term. Both refer to the user sideequipment in a GPON system. GPON supports high-bandwidth, triple-playservices, and distances up to 20 km.

Within a fiber to the business configuration 11714, the OLT 11702 isconnected to the ONU 11704 through an optical fiber 11710. The ONU 11704connects with a PON termination point 11716 via copper wire 11718.Within a fiber to the cabinet configuration 11720, the OLT 11702 isconnected to the ONU 11704 through an optical fiber 11710. The ONU 11704connects with a PON termination point 11716 via copper wire 11718.Finally, within the fiber to the home connection 11722, the OLT 11702connects with the ONT 11706 through a fiber 11710.

Referring now to FIG. 118, a single fiber 11710 from the OLT 11702 runsto a passive optical splitter 11802. The splitter 11802 divides thepower into separate paths 11804 to the users 11806. There can beanywhere between 2 and 128 user paths 11804. GPON has two multiplexingmechanisms. In the downstream direction (OLT to users), data packets areencrypted and broadcast to the users. In the upstream direction (usersto OLT), data packets are transmitted using TDMA.

ONU-ID is an 8-bit identifier that an OLT 11702 assigns to an ONU 11704during ONU activation via PLOAM messages. The ONU-ID is unique acrossthe PON and remains until the ONU 11704 is powered off or deactivated bythe OLT 11702. The OLT 11702 also assigns a 12-bit allocation identifier(ALLOC_ID) to an ONU 11704 to identify a traffic-bearing entity that isa recipient of upstream bandwidth within that ONU.

A transmission container (T-CONT) is a group of logical connections thatappear as a single entity for upstream bandwidth assignment for the ONU11704. The number of supported T-CONTs is fixed for a given ONU 11704.The ONU 11704 autonomously creates all the supported T-CONT instancesduring ONU activation and the OLT 11702 discovers the number of T-CONTinstances supported by a given ONU. There are 5 types of T-CONTs. Type 1is of fixed bandwidth and used for services sensitive to delay and highpriority. Type 2 and 3 are of guaranteed bandwidth types and mainly usedfor video services and data services of high priority. Type 4 is ofbest-effort type and mainly used for data services such as Internet andlow priority. Type 5 is of mixed type involving all bandwidth types.

ONUs 11704 can be located at varying distances from the OLT 11702 whichmeans the transmission delay from each ONU is unique. Ranging isperformed by the OLT 11702 to measure delay and set a register in eachONU 11704 to equalize its delay. The OLT 11702 will transmit a grant toeach ONU 11704 to set a defined interval of time for transmission. Thegrant map is dynamically re-calculated every few milliseconds, and isused to allocate bandwidth to all ONUs for such needs.

Dynamic Bandwidth Allocation (DBA) allows quick adoption of users'bandwidth allocation based on current traffic requirements. GPON usesTDMA for managing upstream access by ONUs 11704, and TDMA providesunshared timeslots to each ONU for upstream transmission. DBA allowsupstream timeslots to shrink and grow based on the distribution ofupstream traffic loads. Without DBA support on the OLT 11702, upstreambandwidth is statically assigned to T-CONTs which cannot be shared andcan be changed only though a management system.

There are two forms of DBA, Status Reporting DBA (SR-DBA) and Non-StatusReporting DBA (NSR-DBA). In SR-DBA the OLT 11702 solicits T-CONT bufferstatus and the ONUs 11704 respond with a separate report for eachT-CONT. The OLT 11702 re-calculates bandwidth allocation once itreceives the report and sends the new map to the ONUs 11704. The ONUs11704 send data in the specified time slots. An ONU 11704 sends idlecell upstream to the OLT 11702 to inform that the ONU has no informationto send. The OLT 11702 can then assign grants to other T-CONTs.

In NSR-DBA an OLT 11702 constantly allocates a small amount of extrabandwidth to each ONU 11704. If the OLT 11702 observes that an ONU 11704is not sending idle frames, it increases the bandwidth allocation forthat ONU. If an ONU 11704 is sending idle frames then the OLT reducesits allocation accordingly. NSR-DBA has the advantage that the ONUs11704 need not be aware of DBA, however the disadvantage is that thereis no way for the OLT 11702 to know how to allocate bandwidth to severalONUs 11704 in the most efficient way.

Referring now to FIG. 119 there are illustrated an Upstream GTS frame11902 and a Downstream GTS frame 11904. FIG. 120 shows a more detailedview of the downstream GTC frame 11904. A downstream GTC frame 11904 hasa duration 11906 of 125 us and is 38880 bytes long which corresponds toa downstream data rate of 2.48832 Gbps. The OLT 11702 broadcasts thePCBd (GTC header) 11908 to every ONU 11704, and the ONUs act upon therelevant information. The PCBd includes the Psync field 12002 thatindicates a beginning of the frame for the ONUs 1704. The Ident field12004 contains an 8-KHz Superframe Counter field. The PLOAMd field 12006handles functions such as OAM-related alarms or threshold-crossingalerts. BIP field 12008 is Bit Interleaved Parity used to estimate biterror rate. The downstream Payload Length indicator (Plend) 12010 givesthe length of the upstream bandwidth (US BW) map. Each entry in theUpstream Bandwidth (US BW) map field 12012 represents a single bandwidthallocation to a particular T-CONT.

The Allocation ID field 12014 indicates the recipient of the bandwidthallocation and uses the lowest 254 allocation ID values to address theONU 11704 directly. The Flag field 12016 allows the upstreamtransmission of physical layer overhead blocks for a designated ONU11704. The Slot Start field 12018 and Stop field 12020 indicates thebeginning and ending of upstream transmission window. The CRC field125022 provides error detection and correction on bandwidth allocationfield.

The GTC payload field 12024 contains a series of GEM (GPON EncapsulationMethod) frames 12026. The downstream GEM frame stream is filtered at theONU 11704. Each ONU 11704 is configured to recognize which Port-IDsbelong to it, and the Port-ID uniquely identifies a GEM Frame 12026.

Referring now back to FIG. 119. The Upstream GTS frame duration 11914 is125 us and is 19440 Bytes long which gives an upstream data rate of1.24416 Gbps. Each frame 11910 contains a number of ONU transmissionbursts 11912 from ONUs 11704 and each burst contains a physical layeroverhead (PLOu) section 11914 and one or more bandwidth allocationintervals 11916. The BW map dictates the arrangement of the burstswithin the frame and the allocation intervals within each burst.

Referring now also to FIG. 121, there is more particularly illustratedan Upstream GTS frame 11910. The PLOu burst 12102 contains the preamblewhich ensures proper physical layer operation. The PLOu field 12102 alsocontains the ONU-ID field 12104 which indicates the unique ONU-ID ofthat ONU 11704. The upstream physical layer OAM (PLOAMu) field 12106 isresponsible for management functions like ranging, activation of an ONT11706, and alarm notifications. The upstream power leveling sequence(PLSu) field 12108 contains information about the laser power levels atthe ONUs 11704 as seen by the OLT 11702.

GEM frames 12026 are transmitted from the OLT 11702 to the ONUs 11704using the GTC frame payload section 12112. The OLT 11702 may allocate upto all of the downstream frame to meet its downstream needs. The ONUfilters the incoming frames based on Port-ID. Frames are transmittedfrom ONU 11704 to OLT 11702 using the configured GEM allocation time.The ONU 11704 buffers the frames and sends them in bursts when allocatedtime by the OLT 11702. The OLT 11702 multiplexes the received framesfrom the ONUs 11704.

Another system that may be used for the optical data transfer systems11504 (FIG. 115) is a NG-PON2 (Next-Generation Passive Optical Network)system. NG-PON2 is a 40 Gbps capable multi-wavelength PON system thatcan grow up to 80 Gbps. An NG-PON2 system has three types of channelrates: basic rate 10/2.5 Gbps or optionally 10/10 Gbps and 2.5/2.5 Gbps.

The main target requirements for NG-PON2 are the increase in aggregatecapacity per Optical Line Terminal (OLT) PON port, a sustainablebandwidth on any Optical Network Unit (ONU) at downstream of 1 Gbps andupstream of 0.5 to 1 Gbps, support of 64 or more ONUs per port,compatibility with legacy PON infrastructure, a 40 km differential reachand a smooth migration, support for multiple applications on the sameOptical Distribution Network (ODN), embedded test and diagnosticscapabilities and PON resilience.

There are many applications driving the demand for next generation PONs,including FTTB, Enterprises, Mobile Backhaul, Fronthaul, and Cloud-RAN.Content is the main driver nowadays for the high access bitraterequirements. Content service providers need to prepare the accessnetwork for the future, and it can be concluded that future accessnetworks will be a truly multi-service solution.

Currently, software packages and personal data is downloaded and storedfrom data centers. This requires very high upload and download speeds aswell as symmetry and low latencies. This means the “cloud opportunity”gained from NG-PON2 is also a very important reason to evolve to the newnetwork.

NG-PON2 is compatible with legacy loss budget classes. NG-PON2 requiresa minimum optical path loss of 14 dB and allows a differential reach of40 km. There are 3 classes defined by NG-PON2 of Tx/Rx wavelengthchannel tuning time. Class 1 components may include switched laser onarrays, class 2 components could be based on electronically tuned lasers(DBR), and class 3 components could be thermally tuned DFBs.

NG-PON2 transmission convergence layer has new capabilities supported bythe protocol, as multiple wavelengths, TWDM and point-to-point channels.Communication starts with a single channel adding more channels laterand distributed OLT Channel Terminations (CTs) which can drive a singlefiber. The new protocol functions allow multiple wavelengths so protocolsupports tuning, new identities to distinguish system and wavelengthchannel, new managements protocol for PtP, WDM and TWDM activation,Dealing with ONUs with uncalibrated lasers that must not be allowed totransmit in the wrong wavelength channel, inter-channel messaging forsome procedures over distributed OLT channel terminations, and new roguescenarios that can be detected and mitigated.

NG-PON2 has an inter-channel termination protocol. The OLT CTs (channeltermination) are distributed so that some procedures require messages tobe passed between OLT CTs such as synchronizing OLT CT quiet windows,ONU tuning, ONU activation, parking orphaned ONUs, ONUs connected to thewrong ODN, guided hand-off of ONUs between OLT CTs, and rogue ONUisolation.

NG-PON2 also covers different protection scenarios and rogue behaviorsof the ONU such as when the ONU transmitter hops to the wrong upstreamchannel, ONU transmitter starts transmitting at the wrong upstreamwavelength, OLT CT transmits in the wrong downstream wavelength channel,and when interference from coexisting devices.

The current NG-PON2 OLT optics are based on Bi-directional OpticalSubassemblies (BOSAs) integrated on XFP form factor. These optics aresuitable for TWDM PON, 10 Gbps downstream, 2.5 Gbps or 10 Gbps upstream.The XFP integrates an electro-absorption integrated laser diode with asemiconductor optical amplifier (SOA) in order to reach the type A N1class NG-PON2 optical requirements. A high sensitivity burst modeavalanche photodiode (APD), a pre-amplifier and a limiting amplifier asreceiver components are mounted into a package integrated in single modefiber-stub with a sensitivity equal to −28.5 dBm at 10 Gbps and −32 dBmat 2.5 Gbps.

The NG-PON2 ONU optics are based on BOSA (Bi-directional Optical SubAssembly) on board. The BOSA integrates a burst mode tunable distributedfeedback lasers (DFB) at 10 Gbps or 2.5 Gbps emitting high optical powerin a N1 type A link, +4-9 dBm capable of doing 4 upstream channels. Onthe receiver side, a high sensitivity 4 channel tunable APD, a preamplifier and a limiting amplifier are able to operate at a sensitivityof −28 dBm at 10 Gbps.

Another implementation that may be used for the optical data transfersystems 11504 (FIG. 115) is a Virtual Optical Line Termination HarwareAbstraction (vOLTHA) that may be used within one of the above describedsystems. As illustrated in FIG. 122, vOLTHA is a layer of abstractionatop legacy and next generation network equipment. It is initially forPON (G-PON, E-PON, XGS-PON) and ultimately for G.Fast, NG-PON2 DOCSIS,and Ethernet. vOLTHA makes an access network act as an abstractprogrammable switch and works with legacy and virtualized devices.vOLTHA can run on the device or on general purpose servers.

Each access technology brings its own protocols and concepts which meanscontrol and management of legacy access devices can be a problem. vOLTHAconfines the differences of access technology to the locality of accessand hides them from the upper layers of the OSS stack. Referring now toFIG. 123 there is illustrated the implementation of vOLTHA with an OLT12302 and ONU 12304 link. The OLT 12302 communicates with multiple ONUs12304 through a splitter 12305. vOLTHA containers communicate over gRPC.The main container publishes events to Kafka and persists data in Consulfor service discovery. Southbound OLT adapters 12306 and ONU adapters12308 will be their own containers as well. OLT adapter 12306 and ONUadapter 12308 enables OLT-ONU interoperability through the vOLTHA core123310. The ONU adapter 12308 sends OMCI (ONT management controlinterface) to the OLT 12302 through the OLT adapter 12306.

Using vOLTHA to create hardware abstraction layers for Wave Agilityenables integration to a residential network IP gateway over mmWaveFixed Wireless Access (gigabit rate access with Dynamic QoS-Application& Network slicing support). One of the challenges faced in the nextgeneration broadband access at gigabit rates is the need for runningfiber to the home or business. Referring now to FIG. 124, with fixedmmWave 5G wireless access technology ONUs 12402 (PON end points) can beutilized for the aggregation of self-installed fixed wireless accesspoints.

Almost all recent FTTH (fiber to the home) deployments, as well as thosecurrently being planned, use passive optical networking. The concept ofa Passive Optical Network (PON) 12412, involves the use of passive fibersplitters which allow multiple customers (typically 32-128) to share asingle fiber pair. GPON has also seen trials and initial deployments byseveral large Telco's, but these are largely used as a basis fortransmitting Ethernet via encapsulation within GEM frames (GPONEncapsulation Method) 12026 (FIG. 120). GPON was designed with verystrict timing requirements. Both EPON and GPON therefore use TimeDivision Multiple Access (TDMA), informally known as “time-sharing.”Time is divided into slots, of either fixed or variable length or longenough to contain one or more data frames (usually around 100-1000msec). During a given slot, one ONU 12402 is permitted to transmit andall others must have turned off their lasers. The OLT 12410 isresponsible for determining a transmission schedule and sending that tothe ONUs 12402 (this is sometimes considered to be a form of batchpolling by the OLT) and the ONUs must maintain an accurate clock whichis synchronized to that of the OLT in order to transmit at exactly theright time.

The number of time slots allocated to each ONU 12402 need not remainfixed. Both EPON and GPON provide flexible mechanisms to allow the OLT12410 to dynamically allocate bandwidth to ONUs according to demand andthe network operator's policy. These mechanisms are nonspecific as tothe algorithms employed, particularly in the case of EPON where theextremely simple request-based protocol leaves a lot of scope forinteresting dynamic bandwidth allocation algorithms. Extending bandwidthassignments to the mmWave technology is desirable with PON technology, achannel is broadcast to all ONUs 12402, and each frame is labelled withthe address of its target ONU. That ONU 12402 will forward the frameonto its end user's LAN through the home gateway 12406, and all otherONUs will discard the frame. This is a form of TDMA, with the OLT 12410determining its own transmission schedule and each time slot lasting theduration of a frame.

A mmWave system 12004 can also take advantage of mmWave beam forming andbeam steering technologies to ensure QoS to the home applicationsaccessed via a home gateway 12406 in the dynamically changing networkconditions. Given the current Residential Gateway (RGW) devices 12406 donot have the ability to directly and dynamically trigger or adjust theservice flow operations based on the network conditions, the hybrid ONU12404 and mmWave Remote Units (RUs) 12408 can be designed withinnovative SDN enabled beam steering mechanisms to achieve high qualityuser experience with dynamic network slicing mechanisms and optimizedOLT-ONU (gPON) signaling frameworks. Millimeter wave frequenciesimplemented by the mmWave system 12404 are roughly defined as bands in24, 28, 39 and 60 GHz. However, such an approach is also applicable to3.5 GHz CBRS. The mmWave system 12404 provides much potential for use aswireless broadband services with beam steering under control of SDNtowards the self-installed mmWave home modems. As mentioned previously,the SDN beam steering mechanisms and dynamic network slicing mechanismsmay use those techniques describe in U.S. patent application Ser. No.15/664,764, which is incorporated herein by reference.

In the vOLTHA scenario, home gateways 12406 can connect to ONUs 12402via mmWave technologies within the mmWave System 12404 in the last drop(100 s of meters) where wireless access points are directly connected toONUs 12402 via mmWave RUs 12408. Hybrid virtual OLT (vOLTHA) 12410 andmmWave Fixed Broadband Wireless technology through the mmWave system12404 can provide self-installed access opportunities to homes andbusinesses. In addition, the synchronous nature of vOLTHA based on gPONcan extend itself to map to beam steering control technology formapping/distribution of ONU traffic among multiple mmWave modems 12408with support for slicing control at home networks. In this scenario, asingle PON 12412 will be seen by an Ethernet switch as a collection ofpoint-to-point links, one per Hybrid ONU 12402+mmWave Radio Unit 12408.The PON 12412 will typically connect up to 128 ONUs 12402 to each OLT12410, and hybrid ONU-RU will connect to multiple mm-wave modemsutilizing beam steering control plans. The mmWave Modems 12408 areself-installed and reduce the need for a fiber connection to thehome/apartments as well as further provide for additional statisticalgain and aggregation points at the ONU+RU at the Ethernet layer,customers served by these PONs 12412 will be on a single large Ethernet.Furthermore, if delay and cost is not a factor, the ONU+RU's areintegrated and can be treated as IP routers with load balancing andslicing capabilities, provide statistical gain and an aggregation point.

Thus, from the operator's perspective, by bridging together all of acentral office's PONs 12412 and serving ONU+RU 12402/12408 at theEthernet layer, customers served by these PONs 12412 will be on a singlelarge Ethernet. Furthermore, if delay and cost is not a factor we canthreat the ONU+RUs 12402/12408 as IP routers with opportunities for loadbalancing and additional slicing capabilities. The system may also bedesigned wherein where transmit is done at higher 60 GHz band channelfrom outside to inside and a and lower 60 GHz band channel from insideto outside.

The current ONUs 12402 in vOLTHA will be complemented with mmWave RUs12408 which will perform beam steering functions with modems installedat each home. In practical scenarios, small cells deployed with each ONU12402 in urban outdoor environments are regularly affected by trees andpassing objects. In millimeter wave beamforming systems, theenvironmental issues such as wind-induced movement, blockage by trees,may be resolved by beam steering technologies under control of SDN whereeach wavelength uses a very narrow beam pattern. The practicalimpairments of a lamppost deployment scenario need be incorporated intothe beamforming system and system design.

Almost all modern PONs 12412 run on Ethernet at some level either usedas the native protocol on an EPON, or encapsulated in GEM on a GPON,with physical and logical topology of a simple Ethernet PON deploymentshown as follows. Ethernet is now predominantly used as a basis for thedata link layer and Internet Protocol (IP) as ubiquitous network layerprotocol. Some networks still use separate fibers for transmission ineach direction (1310 nm and 1490 nm—for bidirectional use). The EthernetPHY is responsible for providing a serialized bit stream facility (only)to the Medium Access Control (MAC) layer. The MAC is responsible fordividing the bit stream into frames. Frames are labelled with a headercontaining, source and destination MAC addresses. This enables thestatistical multiplexing of multiple hosts' frames on a single link.

FIG. 125 illustrates the interface between the ONU 12402 and theplurality of home gateways 12406. A single optical fiber pair 12502provides data to and from the ONU 12402. The ONU 12402 interfaces with amillimeter wave remote unit 12408, having the ability to generate RFbeams 12504 that may be directed toward one or more millimeter waveradio units 12408B associated with a home or business. The interfacebetween the millimeter wave remote units 12408A and 12408B may includeone or more of the building penetration techniques described herein. Themillimeter wave radio units 12408 provide beam steering techniques andslice control techniques enabling the control of the transmission ofdata bidirectionally between the ONU 12402 and home gateways 12406. Themillimeter wave remote units 12408B associated with the home or businessinterface with the home gateways 12406 to provide broadband dataconnections to the associated home or business structure.

Referring now to FIGS. 126 and 127, there are more particularlyillustrated embodiments for broadband data communications between an OLT12410 and devices located within a structure. With respect to FIG. 126,the OLT 12410 is located at a central office/MEC 12602 that may be partof a virtual base band unit (VBBU). The OLT 12410 schedulestransmissions over the fiber 12604 to the ONU's 12402. The OLT 12410connects to a number of ONU's 12402 through optical fiber pairs 12604.The ONU 12402 maintains an accurate clock to sync with the OLT 12410.Associated with the ONU 12402 is a remote unit 12408. The remote unit12408 is part of the millimeter wave system 12404 described hereinabove.The combined ONU/RV is treated as an IP router providing load-balancingand slicing and further providing statistical gain for signaltransmission and acts as an aggregation point for received data. Thecombined ONV/RV also provides for wireless communications with remoteunits associated with structures. The remote unit 12408 is located on alight pole or tower located near a structure and provides the wirelesslast drop connection to a home or business that replaces fiber DSL andcable.

The remote unit 12408 utilizes controlled beamforming and slice controltechniques to generate radio beams 12606 that are transmitted to anexterior millimeter wave transceiver 12608 located on an exterior of thestructure. The exterior millimeter wave transceiver 12608 repeatssignals receive from the exterior hub and allows the signal to penetratethrough the glass or building. The exterior millimeter wave transceiver12608 transmits the broadband data signals through a window or wall126102 and internal millimeter wave transceiver 12612 using one of theabove described techniques for transmitting through a wall or window.The interior millimeter wave transceiver 12612 also utilizes beamformingand slicing techniques as described herein to transmit wireless beams12614 within the structure to a residential gateway 12616. Theresidential gateway 12616 comprises a self-installed home modem thatprovides an interconnection between the broadband data received from theinterior millimeter wave transceiver 12612 and devices located withinthe structure that communicate with the residential gateway 12616 viawired or wireless connections. The OLT 12410, ONU 12402, RU 12408,millimeter wave transceivers 12608/12612 and residential gateway 12616all include a hardware abstraction layer from vOLTHA as previouslydescribed enable a SDN to control the entire end-to-end configuration ofthe components to access the last drop connection.

FIG. 127 illustrates the same structure described with respect to FIG.126 for broadband data transmissions between the OLT 12410 and theinterior millimeter wave transceiver 12612. Rather than illustrating aconnection to a residential gateway 12616, which the system may stilldo, a 60 GHz wireless connection to a pair of virtual reality (VR)goggles 12702 is illustrated. A 60 GHz transceiver dongle 12704, as willbe more fully described herein below, is inserted into a USB port of theinterior millimeter wave transceiver 12612. This provides the abilityfor the interior millimeter wave transceiver 12612 to bidirectionallycommunicate through the 60 GHz transceiver dongle 12704 with the VRgoggles 12702 located on the interior of the structure. The VR goggles12702 may then be used wirelessly with any interior computer or with acentral office without the need for a local computer. While FIG. 127illustrates a 60 GHz wireless link to VR goggles 12702, other types ofdevices may also wirelessly connected to the 60 GHz transceiver dongle12704 in order to enable broadband data transmissions thereto. Thecontrol of data transmissions between the optical data transmissionportions and that the RF data transmission portions using SDN slicing asmentioned hereinabove are applicable to each of the embodiments in FIGS.126 and 127. The OLT 12410, ONU 12402, RU 12408, millimeter wavetransceivers 12608/12612 and VR goggles 12702 all include a hardwareabstraction layer from vOLTHA as previously described enable a SDN tocontrol the entire end-to-end configuration of the components to accessthe last drop connection.

Referring now to FIG. 128, there is illustrated a functional blockdiagram of the 60 GHz transceiver dongle 12704. The 60 GHz transceiverdongle 12704 implements a 60 GHz chipset using for example a Perasotransceiver such as that described hereinabove with respect to FIG. 84B.The chipset implements a low cost, low power, high performanceSuperSpeed USB 3.0 to WiGig device. The chipset includes a USB 2.0 and3.0 device/host system interface 12802. The interface 12802 supportslink speeds up to 2.0 Gbps at 10 m and 1 Gbps at 20 m and it is possibleto configure the chipset as a multi-gigabit WiGig USB adaptor or as the60 GHz wireless connection for a peripheral device through a controlinterface 12804.

The 60 GHz transceiver dongle 12704 incorporates two processors 12806 toprovide the highest flexibility in supporting 801.11ad MACfunctionality. CPU code boot loading is supported from the USB interface12802 or external serial flash 12808. The MAC includes sufficientinternal memory 12810 to buffer data transfers to and from the PHY aswell as receiving/transmitting packets to the host interfaces. Noadditional RAM is required.

The physical layer is capable of modulating/demodulating all control andsingle carrier π/2-BPSK, π/2-QPSK and 16-QAM WiGig coding schemes up toa maximum rate of 4.62 Gbps to a high throughput. LDPC (low densityparity check) forward error correction maximizes performance inunreliable or noisy communication channels. A highly configurableprogrammable IO subsystem is included in the baseband, consisting ofGPIO, UART, SPI, TWI, PWM and JTAG. The firmware incorporates multiplelayers of debug functionality such logging and extensive statistic andevent counters.

The 60 GHz transceiver dongle 12704 may be utilized for many differentapplications including mobile multi-gigabyte wireless connectivity, highquality, low latency wireless UHD 4k displays, wireless dockingstations, I/oh and mobile “sync and go,” small cell backhaul and Wi-Fiinfrastructure and other multi-gigabyte links. The system can be builtto have transmit at a higher band channel from outside to inside andfrom a lower band channel from inside to outside at any center frequency(3.5, 24, 28, 39, 60 GHz).

As shown in FIG. 129, every Ethernet interface is assigned a unique,6-byte MAC address 12902 at the time of manufacture to indicate alocally administered address. This MAC address 12902 includes threebytes 12904 identifying the device's manufacturer using anOrganizationally Unique Identifier (OUI) assigned by the IEEE Societywith the remainder assigned by the manufacturer. It is also possible tooverride the manufacturer-assigned MAC address according to some localscheme. One bit 12906 in the first byte acts as a flag to indicate sucha locally administered address. This bit 12906 is set to zero in everymanufacturer-assigned address. This provides the opportunity to map theONU 12410 to mmWave Radio beams and maintain a table that plays theglue-logic between fixed wireless and OLT/ONU assignment slots

Referring now to FIG. 130, switches 13002 within the optical networks(which may be OLTs, ONVs or ONTs) make use of MAC addresses to bridgetogether multiple point-to-point or shared-medium Ethernet segments13006. When a frame passes through a switch 13002, the switch learns thelocation of the sender. The source address of the frame is stored in aforwarding database 13004 in the switch's memory together with theinterface on which the frame arrived. This is used to direct subsequentframes. The switch 13002 looks for frames' destination addresses in thedatabase 13004 to determine the interface to which the frame should beforwarded. If the switch 13002 has no record of the location of anaddress, the frame can be flooded to all interfaces. This is verywasteful of link capacity and the intention is to prevent this.

MAC addresses can also refer to groups of multiple hosts using anotherflag bit 13008. Currently Ethernet does not natively provide multicastrouting, generally using broadcast for all group addresses, but someswitches 13002 can use a technique known as IGMP (Internet GroupManagement Protocol) snooping to hook into IP multicast and inferEthernet multicast groups.

In summary, the goal is to utilize 5G fixed wireless mmWave and 5G corewith slicing, transport it over vOLTHA and provide similar speeds as Gigpower fiber service (e.g., 1 Gbps) to the home with self-installedmodems. This enables a balancing of data flows between the opticalnetworks associated with the PON 12412 and the RF networks using in oneexample the mmWave System 12404. Assumptions are that our neighborhoodstreet poles are populated with ONUs 12402 plus mmWave Remote Units12408.

It will be appreciated by those skilled in the art having the benefit ofthis disclosure that this regeneration and retransmission of millimeterwaves for building penetration provides a manner for providingmillimeter wave signals inside of a building where the signals do noteffectively penetrate. It should be understood that the drawings anddetailed description herein are to be regarded in an illustrative ratherthan a restrictive manner, and are not intended to be limiting to theparticular forms and examples disclosed. On the contrary, included areany further modifications, changes, rearrangements, substitutions,alternatives, design choices, and embodiments apparent to those ofordinary skill in the art, without departing from the spirit and scopehereof, as defined by the following claims. Thus, it is intended thatthe following claims be interpreted to embrace all such furthermodifications, changes, rearrangements, substitutions, alternatives,design choices, and embodiments.

What is claimed is:
 1. A system for enabling signal penetration into abuilding, comprising: first circuitry, located on an exterior of thebuilding, for transmitting and receiving signals at a first frequencythat experience losses when penetrating into an interior of thebuilding, converting received signals at the first frequency into afirst format that overcome losses caused by penetrating into theinterior of the building over a wireless communications link andconverting received signals in the first format into the signals in thefirst frequency; a first antenna associated with the first circuitry fortransmitting the signals in the first format into the interior of thebuilding via a wireless communications link and for receiving signalsfrom the interior of the building in the first format via the wirelesscommunications link; first power circuitry for providing system power toeach of the first circuitry and the first antenna responsive to aprovided power signal; second circuitry, located on the interior of thebuilding and communicatively linked with the first circuitry via thewireless communications link, for receiving and transmitting theconverted received signals in the first format that counteracts thelosses caused by penetrating into the interior of the building from/tothe first circuitry; a second antenna associated with the secondcircuitry for transmitting the signals in the first format to theexterior of the building via the wireless communications link and forreceiving signals from the exterior of the building in the first formatvia the wireless communications link; second power circuitry forproviding system power to each of the second circuitry and the secondantenna responsive to a power signal; first wireless power transmissioncircuitry located on the interior of the building for generating awireless power signal for transmission to the exterior of the buildingover a wireless power link responsive to the power signal; and secondwireless power transmission circuitry located on the exterior of thebuilding for receiving the wireless power signal over the wireless powerlink and generating the provided power signal responsive to the wirelesspower signal.
 2. The system of claim 1 further comprising: wherein thefirst wireless power transmission circuitry comprises: an interface to apower connection in the interior of the building to connect to the powersignal; a laser transmitter for generating the wireless power signal asa laser signal for transmission on the wireless power link between thefirst and the second wireless power transmission circuitry; and whereinthe second wireless power transmission circuitry comprises aphotovoltaic receiver for receiving the laser signal and generating theprovided power signal responsive thereto.
 3. The system of claim 2,wherein the laser transmitter comprises laser diodes for generating thewireless power signal.
 4. The system of claim 1 further comprising:wherein the first wireless power transmission circuitry comprises: aninterface to a power connection in the interior of the building toconnect to the power signal; and a first inductive coil located on theinterior of the building and connected to receive the power signal;wherein the second wireless power transmission circuitry comprises asecond inductive coil located on the exterior of the building, thesecond inductive coil inductively coupled with the first inductive coilto provide the wireless power link with the first inductive coil; andwherein the second inductive coil provides the provided power signalresponsive to the wireless power signal received on the wireless powerlink.
 5. The system of claim 1 further comprising: wherein the firstwireless power transmission circuitry comprises: an interface to a powerconnection in the interior of the building to connect to the powersignal; and a first magnetic resonator located on the interior of thebuilding and connected to receive the power signal; wherein the secondwireless power transmission circuitry comprises a second magneticresonator located on the exterior of the building, the second magneticresonator magnetically coupled with the first magnetic resonator toprovide the wireless power link with the first magnetic resonator; andwherein the second magnetic resonator provides the provided power signalresponsive to the wireless power signal received on the wireless powerlink.
 6. The system of claim 5 further comprising: a first impedancematching network associated with the first magnetic resonator; and asecond impedance matching network associated with the second magneticresonator.
 7. The system of claim 6, wherein the first and the secondimpedance matching networks overcome losses by controlling a number ofturns in the first and the second magnetic resonators, an area of thefirst and the second magnetic resonators, a permeability of the firstand the second magnetic resonators and a frequency response of the firstand the second magnetic resonators.
 8. The system of claim 1 furthercomprising a rectifying and switching network for converting theprovided power signal from AC to a frequency of operation of the firstand the second wireless power transmission circuitry.
 9. A method forenabling signal penetration into a building, comprising: transmittingand receiving signals at a first frequency that experience losses whenpenetrating into an interior of the building using first circuitrylocated on an exterior of the building; converting received signals atthe first frequency into a first format that overcome losses caused bypenetrating into the interior of the building over a wirelesscommunications link using the first circuitry; converting receivedsignals in the first format into the signals in the first frequencyusing the first circuitry; transmitting the signals in the first formatinto the interior of the building via a wireless communications linkusing a first antenna; receiving signals from the interior of thebuilding in the first format via the wireless communications link usingthe first antenna; powering each of the first circuitry and the firstantenna responsive to a provided power signal using first powercircuitry; receiving and transmitting the converted received signals inthe first format that counteracts the losses caused by penetrating intothe interior of the building from/to the first circuitry via secondcircuitry located on the interior of the building and communicativelylinked with the first circuitry via the wireless communications link;transmitting the signals in the first format to the exterior of thebuilding via the wireless communications link using a second antennaassociated with the second circuitry; receiving signals from theexterior of the building in the first format via the wirelesscommunications link using the second antenna; providing system power toeach of the second circuitry and the second antenna responsive to apower signal using second power circuitry; generating a wireless powersignal for transmission to the exterior of the building over a wirelesspower link responsive to the power signal using first wireless powertransmission circuitry located on the interior of the building; andreceiving the wireless power signal over the wireless power link andgenerating the provided power signal responsive to the wireless powersignal using second wireless power transmission circuitry located on theexterior of the building.
 10. The method of claim 9, wherein the step ofgenerating the wireless power signal comprises: receiving the powersignal via an interface to a power connection in the interior of thebuilding; and generating the wireless power signal as a laser signalusing a laser transmitter for transmission on the wireless power linkbetween the first and the second wireless power transmission circuitryresponsive to the received power signal.
 11. The method of claim 10,wherein step of receiving the wireless power signal comprises receivingthe laser signal and generating the provided power signal responsivethereto using a photovoltaic receiver.
 12. The method of claim 10,wherein the step of generating the wireless power signal comprisesgenerating the wireless power signal using laser diodes.
 13. The methodof claim 9, wherein the step of generating the wireless power signalcomprises: receiving the power signal via an interface to a powerconnection in the interior of the building; inductively coupling asecond inductive coil associated with the second circuitry and thesecond antenna with a first inductive coil associated with the firstcircuitry and the first antenna to provide the wireless power link withthe first inductive coil; and providing the provided power signalresponsive to the wireless power signal received on the wireless powerlink.
 14. The method of claim 9, wherein the step of generating thewireless power signal comprises: receiving the power signal via aninterface to a power connection in the interior of the building;magnetically coupling a second magnetic resonator associated with thesecond circuitry and the second antenna with a first magnetic resonatorassociated with the first circuitry and the first antenna to provide thewireless power link with the first magnetic resonator; and providing theprovided power signal responsive to the wireless power signal receivedon the wireless power link.
 15. The method of claim 9 further comprisingconverting the power signal from AC to a frequency of operation of thefirst and the second wireless power transmission circuitry using arectifying and switching network.
 16. A system for enabling signalpenetration into a building, comprising: first circuitry, located on anexterior of the building, for transmitting and receiving signals at afirst frequency that experience losses when penetrating into an interiorof the building, converting received signals at the first frequency intoa first format that overcome losses caused by penetrating into theinterior of the building over a wireless communications link andconverting received signals in the first format into the signals in thefirst frequency; a first antenna associated with the first circuitry fortransmitting the signals in the first format into the interior of thebuilding via a wireless communications link and for receiving signalsfrom the interior of the building in the first format via the wirelesscommunications link; first power circuitry for providing system power toeach of the first circuitry and the first antenna responsive to aprovided power signal; second circuitry, located on the interior of thebuilding and communicatively linked with the first circuitry via thewireless communications link, for receiving and transmitting theconverted received signals in the first format that counteracts thelosses caused by penetrating into the interior of the building from/tothe first circuitry; a second antenna associated with the secondcircuitry for transmitting the signals in the first format to theexterior of the building via the wireless communications link and forreceiving signals from the exterior of the building in the first formatvia the wireless communications link; second power circuitry forproviding system power to each of the second circuitry and the secondantenna responsive to a power signal; an interface to a power connectionin the interior of the building to connect to the power signal; a lasertransmitter located on the interior of the building for generating alaser power signal for transmission to the exterior of the building overa wireless power link responsive to the power signal; and a photovoltaicreceiver located on the exterior of the building for receiving the laserpower signal over the wireless power link and generating the providedpower signal responsive to the laser power signal.
 17. The system ofclaim 16, wherein the laser transmitter comprises laser diodes forgenerating the laser power signal.
 18. A system for enabling signalpenetration into a building, comprising: first circuitry, located on anexterior of the building, for transmitting and receiving signals at afirst frequency that experience losses when penetrating into an interiorof the building, converting received signals at the first frequency intoa first format that overcome losses caused by penetrating into theinterior of the building over a wireless communications link andconverting received signals in the first format into the signals in thefirst frequency; a first antenna associated with the first circuitry fortransmitting the signals in the first format into the interior of thebuilding via a wireless communications link and for receiving signalsfrom the interior of the building in the first format via the wirelesscommunications link; first power circuitry for providing system power toeach of the first circuitry and the first antenna responsive to aprovided power signal; second circuitry, located on the interior of thebuilding and communicatively linked with the first circuitry via thewireless communications link, for receiving and transmitting theconverted received signals in the first format that counteracts thelosses caused by penetrating into the interior of the building from/tothe first circuitry; a second antenna associated with the secondcircuitry for transmitting the signals in the first format to theexterior of the building via the wireless communications link and forreceiving signals from the exterior of the building in the first formatvia the wireless communications link; second power circuitry forproviding system power to each of the second circuitry and the secondantenna responsive to a power signal; an interface to a power connectionin the interior of the building to connect to the power signal; a firstmagnetic resonator located on the interior of the building and connectedto receive the power signal; a second magnetic resonator located on theexterior of the building, the second magnetic resonator magneticallycoupled with the first magnetic resonator to provide a wireless powerlink with the first magnetic resonator; and wherein the second magneticresonator provides the provided power signal responsive to the powersignal received on the wireless power link.
 19. The system of claim 18further comprising: a first impedance matching network associated withthe first magnetic resonator; and a second impedance matching networkassociated with the second magnetic resonator.
 20. The system of claim19, wherein the first and the second impedance matching networksovercome losses by controlling a number of turns in the first and thesecond magnetic resonator, an area of the first and the second magneticresonator, a permeability of the first and the second magnetic resonatorand a frequency response of the first and second magnetic resonator. 21.The system of claim 18 further comprising a rectifying and switchingnetwork for converting the provided power signal from AC to a frequencyof operation of the first and the second power circuitry.
 22. A systemfor enabling signal penetration into a building, comprising: firstcircuitry, located on an exterior of the building, for transmitting andreceiving signals at a first frequency that experience losses whenpenetrating into an interior of the building, converting receivedsignals at the first frequency into a first format that overcome lossescaused by penetrating into the interior of the building over a wirelesscommunications link and converting received signals in the first formatinto the signals in the first frequency; a first antenna associated withthe first circuitry for transmitting the signals in the first formatinto the interior of the building via a wireless communications link andfor receiving signals from the interior of the building in the firstformat via the wireless communications link; first power circuitry forproviding system power to each of the first circuitry and the firstantenna responsive to a provided power signal; second circuitry, locatedon the interior of the building and communicatively linked with thefirst circuitry via the wireless communications link, for receiving andtransmitting the converted received signals in the first format thatcounteracts the losses caused by penetrating into the interior of thebuilding from/to the first circuitry; a second antenna associated withthe second circuitry for transmitting the signals in the first format tothe exterior of the building via the wireless communications link andfor receiving signals from the exterior of the building in the firstformat via the wireless communications link; second power circuitry forproviding system power to each of the second circuitry and the secondantenna responsive to a power signal; an interface to a power connectionin the interior of the building to connect to the power signal; a firstinductive coil located on the interior of the building and connected toreceive the power signal; a second inductive coil located on theexterior of the building, the second inductive coil inductively coupledwith the first inductive coil to provide a wireless power link with thefirst inductive coil; and wherein the second inductive coil provides theprovided power signal responsive to the power signal received on thewireless power link.